Control device for power conversion device, and motor drive system

ABSTRACT

A control device for a power conversion device according to an embodiment includes a drive control unit controlling a current that is caused to flow through a stator winding of a motor, a drive quantity adjustment unit calculating a drive quantity command value defining a drive quantity of the motor, a magnetic flux observer calculating calculates a first estimated value of a stator magnetic flux of the motor and a first estimated value of a rotor magnetic flux of the motor, a current observer calculating a first estimated value of the current flowing through the stator winding of the motor, and an average correction unit performing calculation on the first estimated values of the stator magnetic flux of the motor. The drive quantity adjustment unit calculates a control quantity defining the drive quantity of the motor on the basis of the first estimated values of the stator magnetic flux of the motor.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromU.S. Provisional Patent Application No. 62/737,125, filed on Sep. 27,2018; the entire contents of which are incorporated herein by reference.

FIELD

Embodiments described herein relate generally to a control device for apower conversion device and a motor drive system.

BACKGROUND

A deadbeat direct torque and flux control (DB-DTFC) system has beenknown as a control system directly controlling a magnetic flux and atorque of a motor. In the DB-DTFC system, a range designated to amagnetic flux of a motor is represented as a circle, and an equation ofa change rate of a requested torque is represented as a straight line ina two-dimensional magnetic flux space. A control device using theDB-DTFC system controls a power conversion device by generating acommand value used for controlling a magnetic flux and a torque of amotor using a relation between a circle and a straight line in amagnetic flux space. In a control device using the DB-DTFC system, it isdesired that an induction motor is controlled with a high accuracy.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating a motor drive system according toa first embodiment;

FIG. 2 is a block diagram illustrating a current/magnetic fluxestimation unit according to the first embodiment;

FIG. 3 is a diagram illustrating a predetermined transformation ruleaccording to the first embodiment;

FIG. 4 is a diagram illustrating advantages of an application of thepredetermined transformation rule based on a synchronous angularfrequency of a motor according to the first embodiment;

FIG. 5 is a timing diagram illustrating DB-DTFC according to the firstembodiment;

FIG. 6A is a diagram illustrating voltage/torque control according tothe first embodiment;

FIG. 6B is a diagram illustrating voltage/torque control according tothe first embodiment;

FIG. 7 is a block diagram illustrating a DB-DTFC calculation unitaccording to the first embodiment;

FIG. 8 is a diagram illustrating voltage/torque control according to afirst modification of the first embodiment;

FIG. 9 is a block diagram illustrating a current/magnetic fluxestimation unit according to a second modification of the firstembodiment;

FIG. 10 is a block diagram illustrating a motor drive system accordingto a second embodiment;

FIG. 11 is a block diagram illustrating a DB-DTFC calculation unitaccording to the second embodiment;

FIG. 12 is a block diagram illustrating a motor drive system accordingto a third embodiment;

FIG. 13 is a block diagram illustrating a control device according to anembodiment;

FIG. 14 is a diagram illustrating a result of evaluation of a motordrive system according to a first example;

FIG. 15 is a diagram illustrating a result of evaluation of a motordrive system according to a second example;

FIG. 16 is a diagram illustrating a result of evaluation of a motordrive system according to a comparative example;

FIG. 17 is a diagram illustrating a result of evaluation of a stepresponse test of torque;

FIG. 18 is a diagram illustrating a result of evaluation of a currentobserver;

FIG. 19 is a diagram illustrating variables of complex vectors amongvariables according to the embodiments;

FIG. 20 is a diagram illustrating variables of scalars among variablesaccording to the embodiments;

FIG. 21 is a diagram illustrating variables of scalars among variablesaccording to the embodiments; and

FIG. 22 is a diagram illustrating advantages of compensation for theamount of lag in calculation of the current observer according to theembodiments.

DETAILED DESCRIPTION

A control device for a power conversion device that drives a motoraccording to an embodiment includes a drive control unit, a drivequantity adjustment unit, a magnetic flux observer, a current observer,and an average correction unit. The drive control unit controls acurrent which the power conversion device causes to flow through astator winding of the motor. The drive quantity adjustment unitcalculates a drive quantity command value defining a drive quantity ofthe motor and controls the power conversion device on the basis of thecalculated drive quantity command value. The magnetic flux observer thatcalculates a first estimated value of a stator magnetic flux of themotor and a first estimated value of a rotor magnetic flux of the motoron the basis of at least the calculated drive quantity command value.The current observer that calculates a first estimated value of thecurrent flowing through the stator winding of the motor on the basis ofat least the calculated first estimated value of the stator magneticflux. The average correction unit that includes a first transformationprocessing unit, a second transformation processing unit, and a thirdtransformation processing unit. The first transformation processing unitcalculates a second estimated value of the stator magnetic flux of themotor for the first estimated value of the stator magnetic flux of themotor in accordance with a first predetermined transformation rule basedon an synchronous/excitation angular frequency of the motor. The secondtransformation processing unit calculates a second estimated value ofthe rotor magnetic flux of the motor for the first estimated value ofthe rotor magnetic flux of the motor in accordance with a secondpredetermined transformation rule based on the synchronous/excitationangular frequency of the motor. The third transformation processing unitcalculates a second estimated value of a stator current of the motor fora first estimated value of the stator current of the motor in accordancewith a third predetermined transformation rule based on thesynchronous/excitation angular frequency of the motor. The drivequantity adjustment unit calculates a control quantity defining thedrive quantity of the motor on the basis of at least a torque commandfor the motor, the second estimated value of the stator magnetic flux ofthe motor, the second estimated value of the rotor magnetic flux of themotor, and the second estimated value of the current flowing through thestator winding of the motor.

Hereinafter, a control device for a power conversion device and a motordrive system according to embodiments will be described with referenceto the drawings. A power conversion device and a motor drive systemdescribed below supply predetermined AC power to a motor.

In the following description, a motor drive system according to anembodiment will be identified as a discrete time system model. Asvariables representing a time history for identifying a calculationcycle, k, (k+1), and (k+2) will be used. A time point in the future byone unit time with respect to a time point k, which is a start point ofa calculation cycle, will be represented as a time point (k+1), and atime point in the future by one more unit time will be represented as atime point (k+2). Any given time point (third time point) between thetime point (k+1) and the time point (k+2) may be represented as a timepoint (k+α). Here, a may be a real number of 1 to 2, and 1.5 is arepresentative value thereof. In the embodiment, in a calculation cycleassigned from a time point (k+1) (first time point) as its start pointto a time point (k+2) (second time point) subsequent to the time point(k+1), an estimated value predicting a state at a time point (k+α) maybe used. A calculation cycle having the time point (k+1) as its startpoint will be referred to as a current cycle, a calculation cycle havingthe time point k as its start point will be referred to as a previouscycle, and a cycle having the time point (k+2) as its start point willbe referred to as a next cycle.

For example, a state quantity based on an operation state of the motor 2in a control cycle switched at the time point k may be sampled at thetime point (k+1), and data of results of this sampling and data ofcommand values or the like generated from the result of the samplingwill be referred to as time history data.

First Embodiment

Next, an example of the configuration of a motor drive system 1 will bedescribed. FIG. 1 is a block diagram illustrating the motor drive system1 according to the embodiment.

The motor drive system 1 includes, for example, a motor 2, a powerconversion device 3, a current detector 9 a, a current detector 9 b, anda control device 10. As illustrated in FIG. 1, the motor drive system 1receives power from an AC power supply 5(G). The control device 10controlling the power conversion device 3 is applied to the motor drivesystem 1.

The motor 2 is, for example, a three-phase induction motor (IM). A shaftof the motor 2 is mechanically coupled with a shaft of a load which isnot illustrated in the drawings. A rotor of the motor 2 rotates by, forexample, three-phase AC power supplied to a stator winding, therebyrotating the shaft of the load. A sensor 2A is disposed on the shaft ofthe motor 2. The sensor 2A includes, for example, a resolver, a velocitysensor, and the like. The sensor 2A detects rotation of the shaft of themotor 2 and outputs an angle (phase) or an angular velocity of theshaft. A torque sensor is not disposed in the motor 2.

The power conversion device 3 includes, for example, a rectifier 6, acapacitor 7, and a power conversion unit 8. The rectifier 6 rectifies analternating current supplied from the AC power supply 5 to the AC inputof the rectifier 6. DC links are connected to DC outputs of therectifier 6. The capacitor 7 is disposed with the DC links. Thecapacitor 7 smoothes a voltage applied to the DC links.

The DC input of the power conversion unit 8 is connected to the DClinks. The power conversion unit 8 converts DC power supplied throughthe DC link into three-phase AC power and supplies the three-phase ACpower from the AC output of the power conversion unit 8 to the motor 2.The power conversion unit 8 is a voltage-type inverter. For example, thepower conversion unit 8 is driven in accordance with pulse widemodulation (PWM) control from the control device 10 to be describedlater. The power conversion unit 8 is controlled by the control device10 with variable voltage variable frequency (VVVF), and adjusts thevelocity and the like of the motor 2.

The power conversion unit 8 includes a power conversion circuitcorresponding to three phases of the AC output. The power conversioncircuit includes an upper arm and a lower arm for each phase. Each ofthe upper arm and the lower arm includes one switching device.

The current detector 9 a is disposed for v phase on the output side ofthe power conversion unit 8. The current detector 9 a detects v-phasestator current Ivs. The current detector 9 b is disposed for w phase onthe output side of the power conversion unit 8. The current detector 9 bdetects w-phase stator current Iws. Although the current detectors 9 aand 9 b illustrated in the drawing are respectively disposed for the twophases, current detectors may be respectively disposed for three phases.

The control device 10 controls the power conversion device 3 on thebasis of command values given by a host device and detection resultsacquired by the current detectors 9 a and 9 b.

Here, a coordinate system used by the control device 10 will bedescribed.

The control executed by the control device 10 uses a plurality ofcoordinate systems, i.e., first to third coordinate systems, accordingto purposes.

The first coordinate system is a three-phase coordinate system. Athree-phase coordinate system includes components of three phases basedon a voltage of the stator winding (stator voltage) of the motor 2. Forexample, the stator voltage of the motor 2 can be represented usingcomponents of three phases including u phase, v phase, and w phase(three-phase signal components). When a stator voltage of the motor 2 isrepresented as a vector on a predetermined plane with respect to anorigin, voltage vectors of the phases have an angular difference of 2π/3therebetween and are drawn radially from the origin (center).

The second coordinate system is a dqs-axis coordinate system. A dqs-axiscoordinate system includes ds axis and qs axis that are orthogonal toeach other. For example, the three-phase coordinate system and thedqs-axis coordinate system may be disposed on a predetermined plane, insuch a manner that, with reference to the origin of the dqs-axiscoordinate system, the direction of the qs axis of the dqs-axiscoordinate system is arranged to match the direction of a voltage vectorof the u phase of the stator. An arithmetic operation of transformingthree-phase signal components of the three-phase coordinate system intotwo-phase signal components of the ds axis and the qs axis of thedqs-axis coordinate system will be referred to as “dqs-axistransformation”. In accordance with the “dqs-axis transformation”,three-phase signal components are transformed into two-phase signalcomponents of the ds axis and the qs axis. An arithmetic operation oftransforming two-phase signal components of the ds axis and the qs axisof the dqs-axis coordinate system into three-phase signal components ofthe three-phase coordinate system will be referred to as “dqs-axisinverse transformation”. In accordance with the “dqs-axis inversetransformation”, the two-phase signal components of the ds axis and theqs axis are transformed into three-phase signal components. For example,the origin of the dqs-axis coordinate system is defined on the basis ofthe stator magnetic flux.

The third coordinate system is a re-aligned coordinate system. Are-aligned coordinate system, similar to the second coordinate system (astator-side coordinate system), includes ds axis and qs axis that areorthogonal to each other. An arithmetic operation of transformingtwo-phase signal components of the ds axis and the qs axis of thestator-side coordinate system into two-phase signal components of the dsaxis and the qs axis of the re-aligned coordinate system will bereferred to as “ras-axis transformation”. In accordance with the“ras-axis transformation”, two-phase signal components of the ds axisand the qs axis of the stator-side coordinate system are transformedinto two-phase signal components of the ds axis and the qs axis of there-aligned coordinate system. An arithmetic operation of transformingtwo-phase signal components of the ds axis and the qs axis of there-aligned coordinate system into two-phase signal components of the dsaxis and the qs axis of the stator-side coordinate system will bereferred to as “ras-axis inverse transformation”. In accordance with the“ras-axis inverse transformation”, two-phase signal components of the dsaxis and the qs axis of the re-aligned coordinate system are transformedinto two-phase signal components of the ds axis and the qs axis of thestator-side coordinate system. The re-aligned coordinate system is usedin DB-DTFC to be described later. In the embodiment, methods of definingthe direction of an axis of the re-aligned coordinate system include twotechniques: a technique using the stator magnetic flux as a reference;and a technique using the rotor magnetic flux as a reference are used.Details of the re-aligned coordinate system will be described later. Forexample, the origin of the re-aligned coordinate system is defined onthe basis of the stator magnetic flux.

In FIGS. 19 to 21, variables used in equations and drawings illustratingembodiments as examples will be described. FIG. 19 is a diagramillustrating, as an example, variables of complex vectors amongvariables according to embodiments. FIGS. 20 and 21 are diagramsillustrating, as an example, variables of scalars among variablesaccording to embodiments.

For example, an estimated value of the stator magnetic flux in thedqs-axis coordinate system is denoted as a stator qds-axis magnetic fluxestimated value λqds_s_est in this embodiment. Here, “λ” represents amagnetic flux. “qds” in a first part of a suffix subsequent theretorepresents a qs-axis component and a ds-axis component of the dqs-axiscoordinates. “s” in a second part of the suffix represents a stationarycoordinate system at the stator side (hereinafter, referred to as astator-side coordinate system). A stator qds-axis magnetic flux λqds_scollectively represents two-phase components of the dqs-axiscoordinates. In the above case, the two-phase components include twocomponents, i.e., a stator qs-axis magnetic flux λqs_s and a statords-axis magnetic flux λds_s. The stator qs-axis magnetic flux λqs_srepresents q-axis component in the stator-side dqs-axis coordinatesystem of the stator magnetic flux. The stator ds-axis magnetic fluxλds_s represents d-axis component in the stator-side dqs-axis coordinatesystem of the stator magnetic flux. In some cases, informationrepresented by two-phase components may be collectively handled as avector value in a complex vector space. “est” of a third part of thesuffix represents an estimated value. Information used for identifyingchronological order information is written within parentheses followingthe third part. There are a command value (com), a differential value(dot), a detection value (det), an average value (ave), and the likeother than those described above that are represented in the third part.

In the following calculation equations and drawings, denotationsdifferent from denotations used in this description may be used. Forexample, the stator qds-axis flux estimated value λqds_s_est may berepresented as in Equation (1).[Math. 1]{circumflex over (λ)}_(qds) ^(s)   (1)

A subscript “qds” of “λ” shown in Equation (1) illustrated aboverepresents information of a two-phase component of the dqs-axiscoordinates. A superscript “s” of “λ” represents information of thestator-side coordinate system. Furthermore, “{circumflex over ( )}”above “λ” represents an estimated value. Other than above, symbols abovea character include “x” representing a differential value. A commandvalue is represented using “*” in a superscript. Variables representingcomplex vectors include the flux magnetic flux λ described above, avoltage V, and a current i. For details of others, please refer to FIGS.19 to 21.

Referring back to FIG. 1, the control device 10 will be described.

The control device 10 includes, for example, a motion controller 12, avelocity/phase estimation unit 13, a DB-DTFC calculation unit 14 (adrive quantity adjustment unit), a first coordinate transformation unit15, a PWM controller 16 (a drive control unit), a second coordinatetransformation unit 17, a slip frequency estimation unit 18, an adderunit 19, a current/magnetic flux estimation unit 20, a delay operationunit 23, a delay operation unit 26, a multiplication unit 27, and anaverage correction unit 30.

The motion controller 12 calculates an air gap torque command valueTe_com(k+1) on the basis of a rotor angular velocity command value(mechanical angle) ωrm_com(k+1) and a rotor angular velocity estimatedvalue (mechanical angle) ωrm_est(k+1). For example, the rotor angularvelocity command value (mechanical angle) ωrm_com(k+1) may be suppliedfrom a device (a host device) external to the control device 10. Therotor angular velocity estimated value (mechanical angle) ωrm_est(k+1)is supplied from the velocity/phase estimation unit 13 to be describedlater. Hereinafter, the rotor angular velocity estimated value(mechanical angle) ωrm_est will be simply referred to as a rotor angularvelocity estimated value ωrm_est. The motion controller 12 calculates anair gap torque command value Te_com(k+1) such that the rotor angularvelocity estimated value ωrm_est(k+1) is caused to follow the rotorangular velocity command value (mechanical angle) ωrm_com(k+1). The airgap torque command value Te_com(k+1) is a command value for the DB-DTFCcalculation unit 14 to be described later.

The velocity/phase estimation unit 13 calculates, for example, the rotorangular velocity estimated value ωrm_est(k+1) and a rotor angleestimated value (electrical angle) θr_est(k+1) on the basis of a rotormechanical angle θrm(k) supplied from the sensor 2A. Hereinafter, therotor angle estimated value (electrical angle) θr_est will be simplyreferred to as a rotor angle estimated value θr_est.

For example, the velocity/phase estimation unit 13 includes a motionobserver that estimates a rotation state of the motor 2. The motionobserver is equivalent to a zero lag filter and decreases a lag of anoutput signal with respect to an input signal such that the lag becomesless than a lag of a generally-used first-order lag filter. In otherwords, the velocity/phase estimation unit 13 decreases lags of the rotorangular velocity estimated value ωrm_est(k+1) and the rotor angleestimated value θr_est(k+1) with respect to the rotor mechanical angleθrm(k). The velocity/phase estimation unit 13 estimates a value of astate quantity at a time point (k+1), i.e., a time point in the futureby one sampling, from a state quantity at a time point in the pastincluding a detection lag of the sensor 2A, for example, at a time pointk. The rotor angular velocity estimated value ωrm_est(k+1) and the rotorangle estimated value θr_est(k+1) respectively serve as estimated valuesof the values of current state quantities. The velocity/phase estimationunit 13 enables acquisition of output signals including low noisecomponents by using such motion observer.

For example, the velocity/phase estimation unit 13 supplies the rotorangular velocity estimated value ωrm_est(k+1) to the motion controller12 and the multiplication unit 27. The rotor angular velocity estimatedvalue ωr_est(k+1) transformed by the multiplication unit 27 is suppliedto the DB-DTFC calculation unit 14, the adder unit 19, and thecurrent/magnetic flux estimation unit 20. The velocity/phase estimationunit 13 supplies the rotor angle estimated value θr_est(k+1) to thedelay operation unit 26. The rotor angle estimated value θr_est(k)delayed by the delay operation unit 26 is supplied to thecurrent/magnetic flux estimation unit 20.

It should be noted that any one of a phase and an angular velocity maybe input to the motion observer described above. When the phase sensoris used as the sensor 2A, the input signal received by thevelocity/phase estimation unit 13 is, for example, a phase θrm(k). Whena velocity sensor is used as the sensor 2A, the input signal received bythe velocity/phase estimation unit 13 is, for example, an angularvelocity ωr(k). A pulse generator (PLG) is an example of a velocitysensor.

When a physical sensor such as the sensor 2A is not used, a positiontracking observer may be used as the velocity/phase estimation unit 13.In this case, as an input signal input to the position tracking observermay be any one of a current, a voltage, and a magnetic flux. For anexample of the configuration of the position tracking observer, refer toYang Xu et al., “Extending Low Speed Self-Sensing via Flux Tracking withVolt-Second Sensing”, [online], 2018, IEEE, [retrieved on Sep. 13,2018], Internet (URL: https://ieeexplore.ieee.org/document/8344841).

The DB-DTFC calculation unit 14 (illustrated as DB-DTFC in the drawing)is a controller that controls the motor 2 in accordance with adeadbeat-direct torque and flux control (DB-DTFC) system. The DB-DTFCcalculation unit 14 calculates a drive quantity command value definingthe drive quantity of the motor 2 on the basis of at least a torquecommand for the motor 2, an estimated value of the stator magnetic fluxof the motor 2, and a reference value of the stator magnetic flux of themotor 2.

For example, the DB-DTFC calculation unit 14 has an air gap torquecommand value Te_com(k+1), a stator qds-axis current estimated valueIqds_s_est(k+α), a stator qds-axis magnetic flux estimated valueλqds_s_est(k+α), a rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α), a stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), a rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1), and a stator qds-axis magnetic flux command valueλqds_s_com(k+2) as input variables and calculates a stator qds-axisvoltage command value Vqds_s_com(k+1) on the basis of the inputvariables described above. The air gap torque command value Te_com(k+1)is supplied from the motion controller 12. The stator qds-axis currentestimated value Iqds_s_est(k+α), the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α), and the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α) are supplied from the average correctionunit 30 to be described later. The stator qds-axis magnetic fluxestimated value λqds_s_est(k+1) and the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+1) are supplied from the current/magneticflux estimation unit 20 to be described later. The stator qds-axismagnetic flux command value λqds_s_com(k+2), for example, may besupplied from a host device or may be calculated in the control device10. The air gap torque command value Te_com(k+1) is an example of atorque command for the motor 2. The stator qds-axis current estimatedvalue Iqds_s_est(k+α) is an example of a second estimated value of thestator current of the motor 2. The stator qds-axis current estimatedvalue Iqds_s_est(k+1) and the stator qds-axis current estimated valueIqds_s_est(k+α) are an example of an estimated value of the statorcurrent of the motor 2. The stator qds-axis magnetic flux estimatedvalue λqds_s_est(k+α) is an example of an estimated value of the statormagnetic flux of the motor 2. The stator qds-axis magnetic fluxestimated value λqds_s_est(k+1) and the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α) are an example of an estimated value ofthe stator magnetic flux of the motor 2. The rotor qds-axis magneticflux estimated value λqdr_s_est(k+α) is an example of an estimated valueof the rotor magnetic flux of the motor 2. The rotor qds-axis magneticflux estimated value λqdr_s_est(k+1) and the rotor qds-axis magneticflux estimated value λqdr_s_est(k+α) are an example of an estimatedvalue of the rotor magnetic flux of the motor 2. The stator qds-axismagnetic flux command value λqds_s_com(k+2) is an example of a referencevalue of the stator magnetic flux of the motor 2. The air gap torquecommand value Te_com(k+1), for example, is acquired at a time point(k+1).

When the DB-DTFC calculation unit 14 calculates the stator qds-axisvoltage command value Vqds_s_com(k+1), the DB-DTFC calculation unit 14performs an RAS coordinate transformation and an inverse RAS coordinatetransformation with reference to the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α). This will be described later.

The DB-DTFC calculation unit 14 outputs the calculated stator qds-axisvoltage command value Vqds_s_com(k+1) to the first coordinatetransformation unit 15 and the delay operation unit 23. The DB-DTFCcalculation unit 14 controls the power conversion device 3 on the basisof the stator qds-axis voltage command value Vqds_s_com(k+1).

The first coordinate transformation unit 15 transforms the statorqds-axis voltage command value Vqds_s_com(k+1) that is a voltage commandvalue in the stator qds-axis coordinate system into three-phase statorvoltage command values Vus_s_com(k+1), Vvs_s_com(k+1), andVws_s_com(k+1) that are voltage command values in the three-phase statorcoordinate system (stationary coordinate system). The transformationexecuted by the first coordinate transformation unit 15 is a “dqs-axisinverse transformation”.

The PWM controller 16 outputs a control signal based on a drive quantitycommand value defining a drive quantity of the motor 2 to the powerconversion device 3 that drives the motor 2. The PWM controller 16compares, for example, the three-phase stator voltage command valuesVus_s com(k+1), Vvs_s_com(k+1), and Vws_s_com(k+1) transformed by thefirst coordinate transformation unit 15 with a carrier signal andgenerates gate pulses GP for the power conversion unit 8 using pulsewidth modulation (PWM). The PWM controller 16 illustrated in FIG. 1outputs the gate pulse GP for each switching device to the switchingdevice of the power conversion unit 8.

The second coordinate transformation unit 17 transforms stator currentsIvs and Iws supplied from the current detectors 9 a and 9 b in discretetimes and transforms the stator currents into a stator qds-axis currentdetection value Iqds_s_det(k) in the stator qds-axis coordinate system.The transformation executed by the second coordinate transformation unit17 is a “dqs-axis transformation”.

The dqs-axis transformation is executed, for example, using thefollowing equation. A stator current Ius is calculated on the basis ofthe stator currents Ivs and Iws. A relationship between three-phasestator currents Ius, Ivs, and Iws and stator currents Iqs_s and Ids_s,which are obtained by the two-phase transformation, is represented inthe following Equation (2). The transformation represented in thefollowing Equation (2) is different from a generally-used Clarketransformation. Please note that the dqs-axis inverse transformation isan inversion of the transformation represented in Equation (2).[Math. 2]Ius+Ivs+Iws=0I _(qs) ^(s) =IusI _(ds) ^(s)=(Ius+2Iws)/√{square root over (3)}  (2)

The slip frequency estimation unit 18 calculates a slip angle frequencyestimated value ωsl_est(k+1) relating to a slip of the motor 2 on thebasis of the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) calculated by a magnetic flux observer 22 to bedescribed later and the stator current detection value Iqds_s_det(k)calculated by the second coordinate transformation unit 17.

The adder unit 19 calculates a synchronous angular frequency estimatedvalue ωe_est(k+1) of the motor 2 by adding the slip angle frequencyestimated value ωsl_est(k+1) calculated by the slip frequency estimationunit 18 to the rotor angular velocity estimated value ωr_est(k+1).

The current/magnetic flux estimation unit 20 includes, for example, acurrent observer 21 and the magnetic flux observer 22.

The current observer 21 calculates an estimated value of a currentflowing through the stator winding of the motor 2 on the basis of atleast a drive quantity command value and an estimated value of thestator magnetic flux. The drive quantity command value is, for example,the stator qds-axis voltage command value Vqds_s_com calculated by theDB-DTFC calculation unit 14.

The current observer 21 outputs an estimated value of the stator currentwith reduced fluctuation on the basis of time history data of theestimated value of the stator magnetic flux calculated by the magneticflux observer 22 to be described later in the calculation processdescribed above. The estimated value of the stator magnetic fluxdescribed above, for example, is a stator qds-axis magnetic fluxestimated value λqds_s_est as chronological order information.

For example, the current observer 21 has the stator qds-axis voltagecommand value Vqds_s_com(k) held by the delay operation unit 23, thestator qds-axis magnetic flux estimated value λqds_s_est(k+1) calculatedby the magnetic flux observer 22, the stator qds-axis current detectionvalue Iqds_s_det(k) transformed by the second coordinate transformationunit 17, and the rotor angular velocity estimated value ωr_est(k+1)calculated by the multiplication unit 27 as input variables andcalculates a stator qds-axis current estimated value Iqds_s_est(k+1) onthe basis of the input variables described above.

The stator qds-axis magnetic flux estimated value λqds_s_est(k+1)described above is an example of a first estimated value of the statormagnetic flux. The stator qds-axis magnetic flux estimated valueλqds_s_est(k) is an example of a previous-cycle estimated value of thestator magnetic flux. The current observer 21 performs a calculationhaving, as an input variable, the stator qds-axis magnetic fluxestimated value λqds_s_est(k) (an estimated value in the past) in aprevious-cycle calculation process corresponding to a time in the pastfrom the calculation process of the current cycle corresponding to thetime point (k+1). The current observer 21 uses the stator qds-axismagnetic flux estimated value λqds_s_est(k) described above for thecalculation process of the current cycle. The current observer 21includes a calculation block 218. The calculation block 218 includes,for example, a zero order hold circuit. Details of the current observer21 will be described later.

The magnetic flux observer 22 calculates at least an estimated value ofthe stator magnetic flux of the motor 2 on the basis of at least thedrive quantity command value calculated by the DB-DTFC calculation unit14 and the output current of the power conversion device 3. The magneticflux observer 22 may use, as one of the variables calculation of theestimated value of the rotor magnetic flux of the motor 2, an estimatedvalue of the current flowing through the stator winding of the motor 2,and calculate the estimated value of the rotor magnetic flux of themotor 2. The magnetic flux observer 22 may calculate an estimated valueof the stator magnetic flux of the motor 2 described above further usingthe rotor angle estimated value θr_est(k) (simply referred to as a rotorangle θr(k)). The magnetic flux observer 22 may calculate at least anestimated value of the stator magnetic flux of the motor 2 further usingthe rotor angle θr(k). The magnetic flux observer 22 calculates anestimated value of the rotor magnetic flux of the motor 2 using theestimated value of the stator magnetic flux of the motor 2. The outputcurrent of the power conversion device 3 described above is a measuredvalue of the output current of the power conversion device 3 and datagenerated on the basis of the output current of the power conversiondevice 3.

For example, the magnetic flux observer 22 has variables including astator qds-axis voltage command value Vqds_s_com(k) held by the delayoperation unit 23, a stator qds-axis current detection valueIqds_s_det(k) transformed by the second coordinate transformation unit17, a stator qds-axis current estimated value Iqds_s_est(k+1) calculatedby the current observer 21, and a rotor angle estimated value θr_est(k),and calculates a stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) on the basis of the input variables described above. Thestator qds-axis voltage command value Vqds_s_com(k) is an example of thedrive quantity command value calculated by the DB-DTFC calculation unit14. The stator qds-axis current estimated value Iqds_s_est(k+1) is anexample of the output current of the power conversion device 3. Thestator qds-axis magnetic flux estimated value λqds_s_est(k+1) is anexample of a first estimated value of the stator magnetic flux of themotor 2. Details of the magnetic flux observer 22 will be describedlater.

The delay operation unit 23 holds the stator qds-axis voltage commandvalue Vqds_s_com(k) calculated by the DB-DTFC calculation unit 14 in astorage unit until calculation of the next cycle. The delay operationunit 23 outputs the stator qds-axis voltage command value Vqds_s_com(k)held in the storage unit to the current/magnetic flux estimation unit 20for a calculation of the current cycle performed by each unit in a laterstage. The stator qds-axis voltage command value Vqds_s_com(k) is anexample of a second value calculated on the basis of the drive quantitycommand value.

The delay operation unit 26 holds the rotor angle estimated valueθr_est(k+1) calculated by the velocity/phase estimation unit 13 in astorage unit until next-cycle calculation. The delay operation unit 26outputs the rotor angle estimated value θr_est(k) that has been held inadvance for a calculation of the current cycle performed by each unit ina later stage.

The multiplication unit 27 calculates a rotor angular velocity estimatedvalue ωr_est(k+1) by multiplying the rotor angular velocity estimatedvalue ωrm_est(k+1) calculated by the velocity/phase estimation unit 13by (P/2) that is the number of pairs of poles. Here, “P” is the numberof poles.

The average correction unit 30 receives input variables including thestator qds-axis current estimated value Iqds_s_est(k+1), the statorqds-axis current estimated value λqds_s_est(k+1), and the rotor qds-axismagnetic flux estimated value λqdr_s_est(k+1) calculated by thecurrent/magnetic flux estimation unit 20, and generates a statorqds-axis current estimated value Iqds_s_est(k+α), the stator qds-axismagnetic flux estimated value λqds_s_est(k+α), and the rotor qds-axismagnetic flux estimated value λqdr_s_est(k+α) corrected on the basis ofthe synchronous angular frequency estimated value ωe_est(k+1) calculatedby the adder unit 19. The average correction unit 30 includes, forexample, a first transformation processing unit 31, a secondtransformation processing unit 32, and a third transformation processingunit 33. A result of calculation executed by the average correction unit30 is used for calculation executed by the DB-DTFC calculation unit 14in a later stage. Details of the average correction unit 30 will bedescribed later.

Here, an overview of the control device 10 according to the embodimentwill be described.

For example, by using the current/magnetic flux estimation unit 20 asbelow, the control device 10 achieves improvement of the accuracy ofestimation of a current and improvement of the accuracy of estimation ofa magnetic flux.

1: The current observer 21 uses an estimated value of a stator magneticflux as input information used for estimating a stator current.

2: The current observer 21 transforms input information of a statormagnetic flux into chronological order information in a discrete timesystem, and estimates a current value corresponding to the inputinformation described above. In this calculation process, the currentobserver 21 performs smoothing of continuous input information which isto be made into the chronological order information.

The current/magnetic flux estimation unit 20 achieves improvement of theaccuracy of estimation of a rotor magnetic flux by further improving theaccuracy of estimation of a stator current. As described above, theDB-DTFC calculation unit 14 calculates a drive quantity command valuedefining a drive quantity of the motor 2 by using at least the estimatedvalue of the stator magnetic flux of the motor 2. Further increasing theaccuracy of estimation of a magnetic flux may contribute to increasingof the accuracy of estimation of a torque calculated on the basis of themagnetic flux. Since the DB-DTFC calculation unit 14 calculates anestimated value of a torque and uses a result thereof for torquecontrol, the accuracy of estimation of the stator current may contributeto improvement of the accuracy of the torque control.

For example, by using information relating to an estimated statequantity of the motor 2 at a time point different from a sampling timepoint relating to the discrete time process as the input information ofthe DB-DTFC calculation unit 14, the control device 10 achievesimprovement of the accuracy of torque estimation and improvement of theaccuracy of a voltage command for the motor 2.

A time point different from the sampling time point in the time axisdirection may be defined, and information at the defined time point isgiven the DB-DTFC calculation unit 14. For example, although informationestimated by the current/magnetic flux estimation unit 20 is informationat the sampling time point described above, the average correction unit30 may correct the information to information of a time point differentfrom the sampling time point in the time axis direction.

Hereinafter, details thereof will be sequentially described.

FIG. 2 is a block diagram illustrating the current/magnetic fluxestimation unit 20 according to the embodiment.

The current/magnetic flux estimation unit 20 includes a current observer21 and a magnetic flux observer 22.

The current observer 21 includes, for example, a calculation block 210,a calculation block 211 (an average calculation unit), a calculationblock 212, a calculation block 213, a calculation block 214, acalculation block 215, a calculation block 216, a calculation block 217,and a calculation block 218.

The calculation block 211 calculates a moving average value using timehistory data of the stator qds-axis magnetic flux estimated valueλqds_s_est, thereby smoothing the stator qds-axis magnetic fluxestimated value λqds_s_est. For example, the calculation block 211 is asampler of the current observer 21 and may derive an average value ofacquired values. In a more specific example, the calculation block 211stores the stator qds-axis magnetic flux estimated value λqds_s_est(k)(a previous-cycle estimated value of the rotor magnetic flux) calculatedby the magnetic flux observer 22 in a previous (k) calculation cycle inthe storage unit. After completion of calculation of an average value tobe described later, the calculation block 211 updates the value storedin the storage unit from the stator qds-axis magnetic flux estimatedvalue λqds_s_est(k) to the stator qds-axis magnetic field estimatedvalue λqds_s_est(k+1). The calculation block 211 temporarily stores thestator qds-axis magnetic field estimated value λqds_s_est(k+1) until itis used in the calculation of the next (k+2) calculation cycle. Thecalculation block 211 uses the stator qds-axis magnetic field estimatedvalue λqds_s_est(k+1) in the calculation of the next (k+2) calculationcycle.

The calculation block 211 calculates a moving-time average valueλqds_s_ave(k) (simply referred to as a moving time average value λave)that is an average of the stator qds-axis magnetic flux estimated valueλqds_s_est(k) and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) on the basis of the stator qds-axis magnetic fluxestimated value λqds_s_est(k) (a previous-cycle estimated value of thestator magnetic flux) and the stator qds-axis magnetic flux estimatedvalue (a first estimated value of the stator magnetic flux) stored inthe storage unit. The stator qds-axis magnetic flux estimated valueλqds_s_est(k) and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) are examples of estimated values of the rotor magneticflux of the motor 2 calculated by the magnetic flux observer 22.

As explained above, it is preferable for the calculation block 211 touse two samples, i.e., the stator qds-axis magnetic flux estimated valueλqds_s_est(k) and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), for the moving average calculation. The reason why thecalculation block 211 uses two samples is to get an accurate estimationof the moving time average value save for the physical system with nodelay. For example, when the stator qds-axis magnetic flux estimatedvalue λqds_s_est(k+1) is used without moving average calculation, thiswill result in phase lead. When the number of samples used for themoving average calculation is more than two samples, this will result inphase lag. Therefore, it is preferable for the calculation block 211 touse two samples for the moving average calculation. When the samplingfrequency fs (fs is a reciprocal of the sampling period t_(s)) issufficiently higher than a fundamental frequency f1 (a synchronousangular frequency ωe), the mathematical model constituted by the currentobserver 21 is accurate. However, when the sampling frequency fs cannotbe made to be sufficiently higher than the fundamental frequency f1, themathematical model constituted by the current observer 21 is inaccurate,and limit cycle is more likely to occur as explained below.

A limit cycle may occur in a discrete time system. The limit cycle is aphenomenon in which a periodical vibration of an output value occurs insynchronization with the sampling frequency fs. The limit cycle tends tohave a larger amplitude as a ratio (fs/f1) of the sampling frequency fsto the fundamental frequency f1 decreases. For example, the ratio(fs/f1) decreases when a calculation period (the sampling period t_(s))increases, or when the fundamental frequency f1 increases.

The calculation block 211 has an effect of suppressing a frequencycomponent of a half of the sampling frequency fs by taking a moving-timeaverage of two consecutive samples. Even when the ratio (fs/f1)described above becomes high, and this limit cycle occurs in a signal ofa sampling target, the calculation block 211 reduces the influence ofnoise having a vibrating property by taking a moving-time average.

The calculation block 212 calculates a voltage correction valueVqds_s_comp1(k) (simply, referred to as a voltage correction valueVcomp1) on the basis of the moving-time average value save and the rotorvelocity ωr. For example, the calculation block 212 calculates a voltagecorrection value Vcomp1 by multiplying the moving-time average valueλave by a transfer function represented in Equation (3) having variablesincluding a rotor resistance Rr, a rotor winding inductance Lr, and arotor velocity ωr.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 3} \right\rbrack & \; \\{\frac{R_{r}}{L_{r}} - {j\;\omega_{r}}} & (3)\end{matrix}$

Equation (3) shown above is an approximation equation for a case where avalue of the rotor velocity ωr is a constant. Alternatively, forexample, the value of the rotor velocity ωr described above may bereplaced with any one of the rotor angular velocity estimated valueωr_est(k), the rotor angular velocity estimated value ωr_est(k+1), avalue acquired by executing a transformation using a predeterminedtransformation rule on the basis of any one of the values describedabove, and a value defined on the basis of values of both the rotorangular velocity estimated value ωr_est(k) and the rotor angularvelocity estimated value ωr_est(k+1). In the above-described case, thevalue of the rotor velocity ωr can be updated.

The calculation block 213 is an adder. The calculation block 213 adds avoltage correction value Vcomp1 calculated by the calculation block 212,the stator qds-axis voltage command value Vqds_s_com, a calculationresult acquired by the calculation block 215, and a calculation resultacquired by the calculation block 216 and outputs the result of additionas a voltage sum value Vqds_s_sum(k+1) (simply referred to as a voltagesum value Vqds_sum).

For example, the calculation block 213 may add at least the voltagecorrection value Vcomp1 calculated by the calculation block 212 and thestator qds-axis voltage command value Vqds_s_com(k) and set a sumthereof as the voltage sum value Vqds_s_tot (simply referred to as avoltage sum value Vtot). The voltage correction value Vcomp1 calculatedby the calculation block 212 is an example of a first value that iscalculated on the basis of a moving-time average value. The statorqds-axis voltage command value Vqds_s_com is an example of a drivequantity command value.

The calculation block 210 is a delay operation circuit. The calculationblock 210 holds an estimation result calculated by the current observer21 in the previous calculation cycle. For example, a value held by thecalculation block 210 is the stator qds-axis current estimated valueIqds_s_est(k).

The calculation block 214 is a subtractor. The calculation block 214subtracts the stator qds-axis current estimated value Iqds_s_est(k) thatis an estimation result of the previous calculation cycle acquired bythe current observer 21 from the stator qds-axis current detection valueIqds_s_det(k) calculated by the second coordinate transformation unit 17and outputs a difference (a deviation ΔIqds) thereof. The calculationblock 215 amplifies the deviation ΔIqds calculated by the calculationblock 214 by multiplying it by a gain K₃. The calculation block 216performs an integration operation with a gain K₄ for the deviation ΔIqdsdescribed above. The calculation block 215 and the calculation block 216form a proportional integration type compensator relating to the currentobserver 21. The calculation block 215 and the calculation block 216generate correction quantities such that the stator qds-axis currentestimated value Iqds_s_est(k) follows the stator qds-axis currentdetection value Iqds_s_det(k). Calculation results acquired by thecalculation block 215 and the calculation block 216 are added in thecalculation block 213 described above.

The calculation block 217 divides the voltage sum value Vqds_sumcalculated by the calculation block 213 by (χσLs) and outputs a quotientthereof. Here, χ is defined as represented in the following Equation(4). σ denotes a leakage coefficient. Ls denotes a stator windinginductance.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 4} \right\rbrack & \; \\{\chi = {\frac{R_{eq}}{\sigma\; L_{s}} - {j\;\omega_{r}}}} & (4)\end{matrix}$

The calculation block 218 has gain characteristics includingcharacteristics of a latch interface. The characteristics of the latchinterface are characteristics of execution of zero-order hold of aninput signal. The calculation block 218 performs zero-order hold of acalculation result acquired by the calculation block 217 and calculatesa stator qds-axis current estimated value Iqds_s_est(k+1) (firstestimated value of a current flowing through a stator winding of themotor) of a current flowing through the stator winding of the motor 2 onthe basis of a result of the zero-order hold. The calculation resultacquired by the calculation block 217 is an example of a valuecalculated on the basis of the moving-time average value λave describedabove. For example, the calculation block 218 calculates a statorqds-axis current estimated value Iqds_s_est(k+1) by multiplying thecalculation result acquired by the calculation block 217 by a transferfunction represented in Equation (5-1) having a variable A1. An exampleof the variable A1 is represented in Equation (5-2).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 5} \right\rbrack & \; \\\frac{1 - A_{1}}{1 - {A_{1}z^{- 1}}} & \left( {5\text{-}1} \right) \\{A_{1} = e^{{- \chi}\; t_{s}}} & \left( {5\text{-}2} \right)\end{matrix}$

The description of the configuration of the current observer 21 will becontinued.

In the following description, a basic range of the current observer 21in which the calculation block 216 is omitted from the calculation block210 and the calculation block 214 will be described.

The current observer 21 uses a latch interface of the calculation block218 for a transformation from a continuous system to a discrete system.This latch interface functions as a zero-order hold function. Since thestator qds-axis voltage command value Vqds_s_com which is a voltageinput to the current observer 21 is based on a command value, thefluctuation of the stator qds-axis voltage command value Vqds_s_comduring a normal operation are relatively small. Meanwhile, the statorqds-axis magnetic field estimated value λqds_s_est of a magnetic fluxinput may include a component of a measured value (a stator qds-axiscurrent measured value Iqds_s_det(k)). For this reason, while thecurrent observer 21 can acquire a relatively accurate value of a voltageinput, a magnetic flux input may be affected by an influence of anexternal disturbance or the like. Thus, for the magnetic flux input, thecurrent observer 21 takes a moving average value of a value acquired inthe current cycle and a value acquired in the previous cycle in order toderive an accurate magnetic flux estimated value without phase lead orlag even at a low fs/f1 ratio.

An equation of a current observer defined in a continuous time system isrepresented in the following Equation (6). Here, “p” denotes adifferential operator.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 6} \right\rbrack & \; \\{{pi}_{qds}^{s} = {\frac{1}{\sigma\; L_{s}}\left\lbrack {V_{qds}^{s} + {\left( {\frac{R_{r}}{L_{r}} - {j\;\omega_{r}}} \right)\lambda_{qds}^{s}} - {R_{eq}i_{qds}^{s}} + {j\;\omega_{r}\sigma\; L_{s}i_{qds}^{s}}} \right\rbrack}} & (6)\end{matrix}$

A sum Vtot of a term of a stator voltage (first term) and a term of astator magnetic flux (second term) in Equation (6) described above isdefined in the following Equation (7). By transforming Equation (7) intoan equation of a discrete time system including average valuecalculation, Equation (7) can be rewritten into Equation (8).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 7} \right\rbrack & \; \\{V_{tot} = {V_{qds}^{s} + {\left( {\frac{R_{r}}{L_{r}} - {j\;\omega_{r}}} \right)\lambda_{qds}^{s}}}} & (7) \\\left\lbrack {{Math}.\mspace{14mu} 8} \right\rbrack & \; \\{{V_{tot}(k)} = {{V_{qds}^{s}(k)} + {\left( {\frac{R_{r}}{L_{r}} - {j\;{\omega_{r}(k)}}} \right)\left( \frac{{\lambda_{qds}^{s}(k)} + {\lambda_{qds}^{s}\left( {k + 1} \right)}}{2} \right)\lambda_{qds}^{s}}}} & (8)\end{matrix}$

By using Equation (7) described above, Equation (6) is rewritten intoEquation (9).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 9} \right\rbrack & \; \\{{pi}_{qds}^{s} = {\frac{1}{\sigma\; L_{s}}\left\lbrack {V_{tot}^{s} - {R_{eq}i_{qds}^{s}} + {j\;\omega_{r}\sigma\; L_{s}i_{qds}^{s}}} \right\rbrack}} & (9)\end{matrix}$

By applying a Laplace transformation to Equation (9), Equation (10) isacquired. “s” in Equation (10) denotes a Laplace operator.[Math. 10](sσL _(s) +R _(eq) −jω _(r) σL _(s))i _(qds) ^(s)(s)=V _(tot) ^(s)(s)+σL_(s) i _(qds) ^(s)(t=0)   (10)

Here, a latch interface having the following Equation (11) as an initialcondition is applied.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 11} \right\rbrack & \; \\{{V_{tot}^{s}(s)} = \frac{V_{tot}^{s}\left( {t = 0} \right)}{s}} & (11)\end{matrix}$

Accordingly, Equation (10) described above is transformed into thefollowing Equation (12).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 12} \right\rbrack & \; \\{{i_{qds}^{s}(s)} = {{\frac{1}{\sigma\; L_{s}}\frac{V_{tot}^{s}\left( {t = 0} \right)}{s\left( {s + \chi} \right)}} + \frac{i_{qds}^{s}\left( {t = 0} \right)}{s + \chi}}} & (12)\end{matrix}$

“χ” in Equation (12) described above is defined as represented in thefollowing Equation (13).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 13} \right\rbrack & \; \\{\chi = {\frac{R_{eq}}{\sigma\; L_{s}} - {j\;\omega_{r}}}} & (13)\end{matrix}$

In accordance with an inverse Laplace transformation, Equation (12) ins-domain is transformed into Equation (14) in time domain.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 14} \right\rbrack & \; \\{{i_{qds}^{s}(t)} = {{\frac{1}{\sigma\; L_{s}}{V_{tot}^{s}\left( {t = 0} \right)}\frac{1}{\chi}\left( {1 - e^{{- \chi}\; t}} \right)} + {{i_{qds}^{s}\left( {t = 0} \right)}e^{{- \chi}\; t}}}} & (14)\end{matrix}$

Equation (14) described above is transformed into Equation (15) of thediscrete time system. This Equation (15) is an example of an equationrepresenting basic characteristics of the current observer 21.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 15} \right\rbrack & \; \\{{{\hat{i}}_{qds}^{s}\left( {k + 1} \right)} = {{\frac{1}{\sigma\; L_{s}}{V_{tot}^{s}(k)}\frac{1}{\chi}\left( {1 - e^{{- \chi}\; t_{s}}} \right)} + {{{\hat{i}}_{qds}^{s}(k)}e^{{- \chi}\; t_{s}}}}} & (15)\end{matrix}$

The calculation block 210, the calculation block 212, the calculationblock 213, the calculation block 214, the calculation block 215, thecalculation block 216, the calculation block 217, and the calculationblock 218 described above are an example of a current estimated valuecalculation unit.

The calculation block 210, the calculation block 212, calculation block213, calculation block 214, calculation block 215, calculation block216, calculation block 217, and calculation block 218 calculate anestimated value (a first estimated value of a current flowing through astator winding of the motor) of a current flowing through the statorwinding of the motor 2 on the basis of a moving-time average value savethat is an average value of the stator magnetic flux of the motor 2.

The calculation block 218 is an example of a smoothing calculation unitthat smooths a continuous output value of the observer described abovein discrete time control. With the calculation block 218, the currentobserver 21 outputs the stator qds-axis current estimated valueIqds_s_est(k+1) smoothed by the calculation block 218. Thecurrent/magnetic flux estimation unit 20 and the current observer 21 arean example of the observer including the calculation block 218.

Since a process of moving average of the stator qds-axis magnetic fluxestimated value λqds_s_est(k+1) is included and executed in anapproximation calculation process of transforming the stator qds-axismagnetic flux estimated value λqds_s_est(k+1) into an average value forthe discrete time model, the current observer 21 can improve theaccuracy of estimation of the stator qds-axis current estimated valueIqds_s_est(k+1) using a simple process. The stator qds-axis magneticflux estimated value λqds_s_est(k+1) is an example of input informationused for estimating the stator qds-axis current estimated valueIqds_s_est(k+1).

Constants that are appropriate for the characteristics of the motor 2are defined as the gain K₃ of the calculation block 215 and the gain K₄of the calculation block 216 illustrated in FIG. 2, so that thecharacteristics of the current observer 21 can be made to be closer tothe characteristics of an actual motor 2, and the influence of externaldisturbances can be decreased. In the case described above, Equation(15) described above may be transformed into an equation including atleast any one of the gain K₃ and the gain K₄. A part or the whole of thecalculation block 216 may be omitted from the calculation block 210 andthe calculation block 214.

Next, the magnetic flux observer 22 will be described. The magnetic fluxobserver 22 illustrated in FIG. 2 includes, for example, a firstmagnetic flux estimation unit 221 and a second magnetic flux estimationunit 222.

The first magnetic flux estimation unit 221 will be described.

The first magnetic flux estimation unit 221 calculates an estimatedvalue of a magnetic flux (a rotor qds-axis magnetic flux estimated valueλqdr_s_cmest(k)) generated by the stator winding of the motor 2 on thebasis of the stator qds-axis current measured value Iqds_s_det(k) andthe rotor angle θr(k), which are measured values, and a calculationequation defined in accordance with a predetermined model. The firstmagnetic flux estimation unit 221 uses a rotary coordinate system havingthe rotor angle θr(k) as a reference phase.

For example, the first magnetic flux estimation unit 221 includes acoordinate transformation block 2211, a calculation block 2212, and acoordinate transformation block 2213.

The coordinate transformation block 2211 transforms a stator qds-axiscurrent measured value Iqds_s_det(k) into a stator qdr-axis currentmeasured value Iqds_r_det(k) that is a variable of the rotary coordinatesystem using a rotor angle θr(k) as a reference phase.

The calculation block 2212 calculates a rotor qdr-axis magnetic fluxestimated value λqdr_r_est(k) by multiplying a transfer functionrepresented in the following Equation (16) by a stator qds-axis currentmeasured value Iqds_r_det(k) calculated by the coordinate transformationblock 2211. The transfer function of the calculation block 2212 isacquired by representing a magnetic flux observer of the motor 2 usingan equation in a discrete time system using a current model of the motor2.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 16} \right\rbrack & \; \\\frac{\begin{matrix}{L_{m}\left\lbrack {\left( {1 - {\tau_{r}\text{/}t_{s}} + {\tau_{r}\text{/}t_{s}e^{{- t_{s}}/\tau_{r}}}} \right) +} \right.} \\\left. {\left( {{\tau_{r}\text{/}t_{s}} - {\tau_{r}\text{/}t_{s}e^{{- t_{s}}/\tau_{r}}} - e^{{- t_{s}}/\tau_{r}}} \right)z^{- 1}} \right\rbrack\end{matrix}}{1 - {\left( e^{{- t_{s}}/\tau_{r}} \right)z^{- 1}}} & (16)\end{matrix}$

The coordinate transformation block 2213 transforms the rotor qdr-axismagnetic flux estimated value λqdr_r_est(k) calculated by thecalculation block 2212 into a magnetic flux estimated value of avariable of a stationary coordinate system using the rotor angle θr(k)as a reference phase. A magnetic flux estimated value that is a resultof this calculation is calculated on the basis of a current model. Thecoordinate transformation block 2213 acquires a rotor qds-axis magneticflux estimated value λqdr_s_cmest(k) through the transformationdescribed above.

The second magnetic flux estimation unit 222 will be described.

The second magnetic flux estimation unit 222 calculates an estimatedvalue of the magnetic flux generated by the stator winding of the motor2 (a stator qds-axis magnetic flux estimated value λqds_s_est(k)) and arotor qds-axis magnetic flux estimated value λqdr_s_est(k) on the basisof the stator qds-axis current measured value Iqds_s_det(k), the statorqds-axis voltage command value Vqds_s_com(k), the rotor qds-axismagnetic flux estimated value λqdr_s_cmest(k), and the stator qds-axiscurrent estimated value Iqds_s_est(k+1). The second magnetic fluxestimation unit 222 uses a calculation equation defined in accordancewith a voltage model of the motor 2 for the calculation described above.

The second magnetic flux estimation unit 222 includes, for example, acalculation block 2221, a calculation block 2222, a calculation block2223, a calculation block 2224, a calculation block 2225, a calculationblock 2226, a calculation block 2227, a calculation block 2228, acalculation block 2229, a calculation block 2230, and a calculationblock 2231.

The calculation block 2221 calculates a stator qds-axis voltage measuredvalue Vqds_s_det(k) corresponding to the stator qds-axis currentmeasured value Iqds_s_det(k) by multiplying the stator qds-axis currentmeasured value Iqds_s_det(k), which is a measured value, by the statorresistance Rs.

The calculation block 2222 is a subtractor. The calculation block 2222subtracts the stator qds-axis voltage measured value Vqds_s_det(k),which is a calculation result acquired by the calculation block 2221,from the stator qds-axis voltage command value Vqds_s_com(k), andoutputs a deviation ΔVqds1 that is a difference therebetween.

The calculation block 2223 is a subtractor. The calculation block 2223subtracts the rotor qdr-axis magnetic flux estimated value λqdr_s_est(k)from the rotor qds-axis magnetic flux estimated value λqdr_s_cmest(k),and outputs a deviation Δλqds2 that is a difference therebetween. Thecalculation block 2224 amplifies the deviation Δλqds2 calculated by thecalculation block 2223 by multiplying it by a gain K_(p). Thecalculation block 2225 performs an integration operation with a gainK_(i) for the deviation Δλqds2 described above. The calculation block2224 and the calculation block 2225 a a compensator of a proportionalintegration type relating to the current observer 21. The calculationblock 2224 and the calculation block 2225 generate a correction quantityfor causing the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k) to follow the rotor qds-axis magnetic flux estimated valueλqdr_s_cmest(k). The calculation results acquired by the calculationblock 2224 and the calculation block 2225 are added by the calculationblock 2226 described above.

The calculation block 2226 is an adder. The calculation block 2226 addsthe deviation ΔVqds1 calculated by the calculation block 2222, thecalculation result acquired by the calculation block 2224, and thecalculation result acquired by the calculation block 2225, and outputsthe result of addition as a voltage adjustment value Vqds_vsum.

The calculation block 2227 calculates a stator qds-axis magnetic fluxestimated value λqds_s_est(k+1) by integrating the voltage adjustmentvalue Vqds_vsum calculated by the calculation block 2226.

The calculation block 2228 multiplies the stator qds-axis currentestimated value Iqds_s_est(k+1) calculated by the current observer 21 bya gain (σLs).

The calculation block 2229 is a subtractor. The calculation block 2229subtracts a calculation result acquired by the calculation block 2228from the stator qds-axis magnetic flux estimated value λqds_s_est(k+1),and outputs a rotor magnetic flux adjustment value that is a differencetherebetween.

The calculation block 2230 multiplies the rotor magnetic flux adjustmentvalue, which is a calculation result acquired by the calculation block2229, by (Lr/Lm) and outputs a rotor qds-axis magnetic flux estimatedvalue λqdr_s_est(k+1). Lm denotes magnetizing inductance.

The calculation block 2231 is a delay operation circuit. The calculationblock 2231 holds the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k) that is a estimation result calculated by the calculationblock 2230 in a previous calculation cycle. The calculation block 2231maintains the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) in the current calculation cycle.

For detailed description of the magnetic flux observer 22, for example,please refer to literatures such as R. D. Lorenz, “The Emerging Role ofDead-beat, Direct Torque and Flux Control in the Future of IndicationMachine Drives”, [online], 2008, IEEE, [retrieved on Sep. 13, 2018],Internet (URL: https://ieeexplore.ieee.org/document/4602331/).

Since the current/magnetic flux estimation unit 20 includes thecombination of the current observer 21 and the magnetic flux observer22, the current/magnetic flux estimation unit 20 achieve characteristicsappropriate for the characteristics of the motor 2. The current observer21 of the current/magnetic flux estimation unit 20 uses the statorqds-axis magnetic field estimated value λqds_s_est(k+1) as an inputvariable. As a result, the interference with the magnetic flux observer22 can be alleviated.

An average correction using the average correction unit 30 will bedescribed with reference to FIG. 1 described above.

The first transformation processing unit 31 of the average correctionunit 30 calculates the stator qds-axis magnetic flux estimated valueλqds_s_est(k+α) (a second estimated value of the stator magnetic flux ofthe motor 2) for the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) (a first estimated value of the stator magnetic flux ofthe motor 2) using a first predetermined transformation rule based onthe synchronous angular frequency ωe of the motor 2.

The second transformation processing unit 32 calculates the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+α) (a secondestimated value of the rotor magnetic flux of the motor 2) for the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1) (a firstestimated value of the rotor magnetic flux of the motor 2) in accordancewith a second predetermined transformation rule based on the synchronousangular frequency ωe of the motor 2.

The third transformation processing unit 33 calculates a stator qds-axiscurrent estimated value Iqds_s_est(k+α) (a second estimated value of thestator current of the motor 2) for the stator qds-axis current estimatedvalue Iqds_s_est(k+1) (a first estimated value of the stator current ofthe motor 2) in accordance with a third predetermined transformationrule based on the synchronous angular frequency ωe of the motor 2. Itshould be noted that the first predetermined transformation rule, thesecond predetermined transformation rule, and the third predeterminedtransformation rule may be the same as each other, or may be differentfrom each other.

The average correction unit 30 described above improves theapproximation accuracy when an equation is transformed from a continuoussystem to a discrete time system using an estimated value of a timepoint that cannot be generated in a period of a calculation cycle byperforming a calculation process according to the predeterminedtransformation rule to be described next. The estimated value describedabove is calculated in accordance with the predetermined transformationrule based on the synchronous angular frequency ωe. The averagecorrection unit 30 adjusts a correction coefficient according to thepredetermined transformation rule in accordance with the magnitude ofthe synchronous angular frequency ωe of the motor 2.

For example, the average correction unit 30 advances the phases of thestator qds-axis magnetic flux estimated value λqds_s_est(k+1), the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1), and the statorqds-axis current estimated value Iqds_s_est(k+1) by the predeterminedangle explained above to yield the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α), the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α), and a stator qds-axis current estimatedvalue Iqds_s_est(k+α). In this way, by performing calculation ofadjusting a phase of a fixed quantity, each signal described above canbe generated.

A predetermined angle corresponding to the synchronous angular frequencyωe of the motor 2 described above is defined by a period in which thedrive quantity adjustment unit calculates a drive quantity command valueand the synchronous angular frequency ωe. The sampling period t_(s) isan example of the period in which the drive quantity adjustment unitcalculates a drive quantity command value.

The synchronous angular frequency ωe of the motor 2 described above isan example of an estimated value of a control state of the motor. Theaverage correction unit 30 calculates a feedback quantity of feedbackcontrol for the DB-DTFC calculation unit 14 by adjusting the statorqds-axis magnetic flux estimated value λqds_s_est(k+1), the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1), and the statorqds-axis current estimated value Iqds_s_est(k+1) on the basis of theestimated value of the control state of the motor 2.

The predetermined transformation rule based on the synchronous angularfrequency ωe according to the embodiment will be described withreference to FIGS. 3 and 4. FIG. 3 is a diagram illustrating thepredetermined transformation rule according to the embodiment. In a dqscoordinate system illustrated in the drawing, a ds axis directeddownward in the drawing and a qs axis that is orthogonal to the ds axisare illustrated. The qs axis is located at a position rotated by (2/π)(radian) in the counterclockwise direction from the ds axis withreference to an intersection between the ds axis and the qs axis, thatis, the origin of the dqs coordinate system. A plurality of arrowshaving the origin of the dqs coordinate system as start points thereofare illustrated. Each of these arrows is a complex vector representing amagnitude and a direction of the magnetic flux λ in the dqs coordinatesystem. For example, when the starting point (tail) of the complexvector of the magnetic flux is disposed at the origin of the dqscoordinate system, for example, the magnetic flux rotates in thecounterclockwise direction about the origin of the dqs coordinate systemin order the following order: a magnetic flux λ(k+1), a magnetic fluxλ(k+1.5), and a magnetic flux λ(k+2). The magnetic flux λ(k+1)illustrated in the drawing is an example of the stator qds-axis magneticflux estimated value λqds_s_est(k+1), the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+1), and the stator qds-axis currentestimated value Iqds_s_est(k+1).

By rotating the magnetic flux λ(k+1) as a start point around the originby (ωe×ts), it reaches the magnetic flux λ(k+2). For example, byrotating the magnetic flux λ(k+1) as a start point by (ωe×ts)/2, itreaches the magnetic flux λ(k+1.5). It is expected for the magnetic fluxλ(k+1.5) to take a value close to an average of the magnetic flux λ(k+1)and the magnetic flux λ(k+2). The description presented above is a basisof the predetermined transformation rule based on the synchronousangular frequency ωe of the motor 2. In the above explanation, forexample, the starting point (tail) of the complex vector of the magneticflux is disposed at the origin of the dqs coordinate system. However,the present embodiment is not limited thereto. Alternatively, thepointing end (tip) of the complex vector of the magnetic flux may bealigned with the origin of the dqs coordinate system.

Meanwhile, although there is a calculation cycle having the time point(k+1) as a start point in a period from the time point (k+1) to the timepoint (k+2), there is no calculation cycle of which start point is atime point between the time point (k+1) and the time point (k+2). Thus,in the embodiment, the magnetic flux λ(k+1.5) is estimated on the basisof the magnetic flux λ(k+1). The magnetic flux λ(k+1.5) can be derivesby rotating the magnetic flux λ(k+1) by a predetermined angle in arotating direction of the rotor.

A transformation for rotating the magnetic flux λ(k+1) by (ωe×ts)/2 isrepresented in the following Equation (17).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 17} \right\rbrack & \; \\\begin{matrix}{{\lambda\left( {k + 1.5} \right)} = {\int_{0}^{{ts}/2}{{\lambda\left( {k + 1} \right)}e^{j\;\omega_{e}t}{dt}}}} \\{= {{\lambda\left( {k + 1} \right)}{\int_{0}^{{ts}/2}{e^{j\;\omega_{e}t}{dt}}}}} \\{= {{\lambda\left( {k + 1} \right)}\frac{1}{j\;\omega_{c}\frac{ts}{2}}e^{j\;\omega_{e}\frac{ts}{2}}}} \\{= {{Ke}\;{\lambda\left( {k + 1} \right)}}}\end{matrix} & (17)\end{matrix}$

Ke represented in Equation (17) described above is defined in thefollowing Equation (18).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 18} \right\rbrack & \; \\{{Ke} = {\frac{2}{\omega_{e}t_{s}}\sin\mspace{11mu}\left( \frac{\omega_{e}t_{s}}{2} \right)e^{\frac{j\;\omega_{e}t_{s}}{2}}}} & (18)\end{matrix}$

FIG. 4 is a diagram illustrating advantages of an application of thepredetermined transformation rule based on the synchronous angularfrequency ωe of the motor 2 according to the embodiment. In a graphillustrated in FIG. 4, an air gap torque (Nm) corresponding to an elapseof time (seconds) is represented.

In the DB-DTFC, a torque change rate during one sampling is necessaryfor torque control. Here, a method of calculating a change rate of thetorque on the basis of an actual air gap torque Te_act will bedescribed.

Although an actual air gap torque Te_act can be measured using a torquesensor, the cost increases and the measured torque has a delay of onesampling period. For this reason, instead of the value measured with thetorque sensor, the actual air gap torque Te_act may also use anestimated value at the time point (k+1) as a value at the time point(k+1) and a command value at the time point (k+1) that is a commandvalue for commanding a value at the time point (k+2) as a value at thetime point (k+2).

The reason for this is that a torque response follows a torque referencein one sampling in accordance with deadbeat control. A torque at thetime point (k+1) will be indicated by the air gap torque estimated valueTe_est(k+1), and a torque at the time point (k+2) will be indicated bythe air gap torque command value Te_com(k+1).

While the time elapses from the time point (k+1) to the time point(k+2), the air gap torque Te changes from the actual air gap torqueTe_act(k+1) to the air gap torque Te_act(k+2). For the simplification ofdescription, it is assumed that the actual air gap torque Te_act(k+1) atthe time point (k+1) coincides with the air gap torque estimated valueTe_est(k+1), and the air gap torque Te at the time point (k+2) coincideswith the air gap torque command value Te_cmd(k+1). A curve Te(k)illustrated in FIG. 4 represents the actual air gap torque Te_actassumed at the time point (k+1).

In the embodiment, two solutions including the first and secondsolutions are given as techniques for acquiring a change rate of the airgap torque Te. In any of the two solutions, a change rate (Te_dot(k+1))of the air gap torque Te at the time point (k+1) is approximated as achange rate (ΔTe_est(k+1)/ts) per unit time of the air gap torque Te.

In the first solution, a change rate of the air gap torque Te in asection from the time point (k+1) to the time point (k+2) is calculatedusing a state quantity at the time point (k+1). In the first solutiondescribed above, a change rate (Te_dot(k+1)) of the air gap torque Te atthe time point (k+1) is defined as in the following Equation (19).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 19} \right\rbrack & \; \\{{{{\overset{.}{T}}_{e}\left( {k + 1} \right)} \approx \frac{\Delta\;{{\hat{T}}_{e}\left( {k + 1} \right)}}{t_{s}}} = {\frac{{T_{e}^{*}\left( {k + 1} \right)} - {{\hat{T}}_{e}\left( {k + 1} \right)}}{t_{s}} = {{\frac{3\mspace{11mu}{PL}_{m}}{4\mspace{11mu}{\sigma L}_{s}L_{r}}\left\lbrack {{{V_{qs}\left( {k + 1} \right)}{\lambda_{dr}\left( {k + 1} \right)}} - {{V_{ds}\left( {k + 1} \right)}{\lambda_{qr}\left( {k + 1} \right)}} - {{\omega_{r}\left( {k + 1} \right)}\left( {{{\lambda_{qs}\left( {k + 1} \right)}{\lambda_{qr}\left( {k + 1} \right)}} + {{\lambda_{ds}\left( {k + 1} \right)}{\lambda_{dr}\left( {k + 1} \right)}}} \right)}} \right\rbrack} - {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right){{\hat{T}}_{e}\left( {k + 1} \right)}}}}} & (19)\end{matrix}$

In the case of the first solution, as represented in Equation (19)described above, a change rate (Te_dot(k+1)) of the air gap torque Te isdefined as a function having variables including the air gap torqueestimated value Te_est(k+1), the air gap torque command valueTe_com(k+1), and the sampling period ts. Since the sampling period is isa constant, the equation described above can be defined withoutdepending on the status of the change in the actual air gap torqueTe_act. In the case of Equation (19) described above, the change rate ofthe air gap torque Te is calculated from a state quantity at the timepoint (k+1). There is an approximation technique of Euler relating tothe description presented above. In the approximation technique ofEuler, an air gap torque estimated value Te_est(k+2) is calculated froma slope of the air gap torque Te at the time point (k+1). The firstsolution is different from a so-called approximation technique of Euler.

In the second solution, a change rate of the torque in a section of thetime point (k+1) to the time point (k+2) is calculated using a statequantity of a time point (k+α).

For example, when a torque in the section from the time point (k+1) tothe time point (k+2) linearly increases as represented as a linearfunction with respect to an elapse of time, a line representing thetorque is a curved line. In such a case, it is preferable that a changerate of the torque in a section from the time point (k+1) to the timepoint (k+2) is calculated as a change rate at the time point (k+1.5)which is a central value thereof. Hereinafter, (k+1.5) will be referredto as (1+α).

During a control cycle from the time point (k+1) to the time point(k+2), a voltage output by the power conversion device 3 is constant. Achange rate of the rotation velocity during the control period is muchsmaller than a change rate of the current or the magnetic flux.Accordingly, for the voltage and the rotation velocity, a state quantityat the time point (k+1) is used for calculation of a change rate of thetorque in the section from the time point (k+1) to the time point (k+2).

Here, the waveforms of the magnetic field and the current are sinusoidalwaves, and instant values thereof change in accordance with elapse oftime. In a case in which a change rate of the torque in the section fromthe time point (k+1) to the time point (k+2) is approximated using statequantities of the magnetic flux and the current at the time point (k+1),approximation error included in the change rate of the torque increasesbecause the values will change after the time point (k+1).

Thus, for the magnetic flux and the current, instead of using the valuesat the time point (k+1), an estimated value at a time point (k+α) isused. This is schematically illustrated in FIG. 4. A straight line Tsol1is a line that represents a change rate of the torque approximated usingthe state quantity at the time point (k+1). A straight line Tsol2 is aline that represents a change rate of the torque approximated using thestate quantity at the time point (k+α).

In the second solution described above, the change rate of the air gaptorque Te at the time point (k+α) is defined as represented in thefollowing Equation (20).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 20} \right\rbrack & \; \\{{{{\overset{.}{T}}_{e}\left( {k + 1} \right)} \approx \frac{\Delta\;{{\hat{T}}_{e}\left( {k + 1} \right)}}{t_{s}}} = {\frac{{T_{e}^{*}\left( {k + 1} \right)} - {{\hat{T}}_{e}\left( {k + 1} \right)}}{t_{s}} = {{\frac{3\mspace{11mu}{PL}_{m}}{4\mspace{11mu}{\sigma L}_{s}L_{r}}\left\lbrack {{{V_{qs}\left( {k + 1} \right)}{\lambda_{dr}\left( {k + \alpha} \right)}} - {{V_{ds}\left( {k + 1} \right)}{\lambda_{qr}\left( {k + \alpha} \right)}} - {{\omega_{r}\left( {k + 1} \right)}\left( {{{\lambda_{qs}\left( {k + \alpha} \right)}{\lambda_{qr}\left( {k + \alpha} \right)}} + {{\lambda_{ds}\left( {k + \alpha} \right)}{\lambda_{dr}\left( {k + \alpha} \right)}}} \right)}} \right\rbrack} - {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right){{\hat{T}}_{c}\left( {k + 1} \right)}}}}} & (20)\end{matrix}$

In the following embodiment, the second solution described above will beillustrated as an example, and an operation thereof will be described.

FIG. 5 is a timing diagram illustrating DB-DTFC according to theembodiment. In the timing diagram illustrated in the drawing, time(seconds) is assigned to the horizontal axis, steps are divided intofirst to third steps in the vertical axis, and a process of each step isrepresented in the vertical axis. Times k, (k+1), and (k+2) respectivelyrepresent times at start points of calculation cycles in discrete timecontrol.

In this timing diagram illustrated in FIG. 5, the first step (STEP 1) inthe upper row includes a sampling process, the second step (STEP 2) inthe middle row includes a calculation process, and the third step (STEP3) in the lower row includes an output control process.

Hereinafter, a calculation cycle of which start point is a time point(k+1) will be described as an example.

In a sampling process of the first step (STEP1), the velocity/phaseestimation unit 13 acquires rotor mechanical angle θrm. For example, therotor mechanical angle θrm is detected using a sensor 2A. The secondcoordinate transformation unit 17 generates a stator qds-axis currentdetection value Iqds_s_det(k) on the basis of v-phase stator current Ivsand w-phase stator current Iws of the motor 2 operating in accordancewith a command value of a calculation cycle of which start point is atime point k. The v-phase stator current Ivs and the w-phase statorcurrent Iws are respectively acquired from the current detectors 9 a and9 b. The rotor mechanical angle θrm, the v-phase stator current Ivs, thew-phase stator current Iws, and the stator qds-axis current detectionvalue Iqds_s_det(k) are examples of state quantities representing thecontrol state of the motor 2.

In the first step described above, the control device 10 acquires astate quantity representing at least a control state of the motor 2. Thecontrol device 10 may acquire an air gap torque Te(k) and may calculatean air gap torque Te(k+1) on the basis of a rotor angular velocitycommand value (mechanical angle) ωrm_com(k+1).

In the second step (STEP2), the process is executed on the basis of aresult of the sampling process executed in the first step. In the secondstep, calculation processes of the current/magnetic flux estimation unit20, the average correction unit 30, the DB-DTFC calculation unit 14, andthe first coordinate transformation unit 15 are executed, and a statorqds-axis voltage command value Vqds_s_com(k+1) that is a drive quantitycommand value for the motor 2 is calculated.

First, as described above, the current/magnetic flux estimation unit 20calculates the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), the stator qds-axis current estimated valueIqds_s_est(k+1), and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1). During the calculation cycle, first, thecurrent/magnetic flux estimation unit 20 calculates the stator qds-axismagnetic flux estimated value λqds_s_est(k+1), subsequently calculatesthe stator qds-axis current estimated value Iqds_s_est(k+1), andthereafter calculates the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1).

Next, the average correction unit 30 calculates the stator qds-axismagnetic flux estimated value λqds_s_est(k+α) the stator qds-axiscurrent estimated value Iqds_s_est(k+α), and the rotor qds-axis magneticflux estimated value λqdr_s_est(k+α) on the basis of the stator qds-axismagnetic flux estimated value λqds_s_est(k+1), the stator qds-axiscurrent estimated value Iqds_s_est(k+1), and the rotor qds-axis magneticflux estimated value λqdr_s_est(k+1).

Next, the DB-DTFC calculation unit 14 calculates ΔTe_est(k+1) and thestator qds-axis voltage command value Vqds_s_com(k+1) using the inputvariables including at least the stator qds-axis magnetic flux estimatedvalue λqds_s_est(k+α), the stator qds-axis current estimated valueIqds_s_est(k+α), and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α).

Next, the first coordinate transformation unit 15 generates athree-phase stator voltage command value Vuvws_com(k+1) that is avoltage reference by performing a dqs-axis inverse transformation forthe stator qds-axis voltage command value Vqds_s_com(k+1).

In the second step described above, a drive quantity command value iscalculated by the control device 10 on the basis of at least the statequantities acquired in the first step, an estimated value of the statormagnetic flux of the motor 2, an estimated value of a current flowingthrough the stator winding of the motor 2, and an estimated value of therotor magnetic flux of the motor 2.

In the third step (STEP3), an output process of supplying a controlsignal based on the drive quantity command value to the power conversiondevice is executed by the control device 10.

The third step is executed on the basis of the detection resultsacquired in the second step. At a calculation start time point at thetime point (k+1), the PWM controller 16 outputs a gate pulse Duty(k) onthe basis of a result of comparison between a three-phase stator voltagecommand value Vuvws_com(k) and carrier signal. At a calculation starttime point at the time point (k+2), the PWM controller 16 updates thethree-phase stator voltage command value with Vuvws_com(k+1) andgenerates a gate pulse GP of Duty(k+1) on the basis of comparison withthe carrier signal. Accordingly, the PWM controller 16 supplies the gatepulse GP of Duty(k+1) to the power conversion device 3.

Also after the time point (k+2), a process similar to that of thecalculation cycle of the time point (k+1) is repeated.

Voltage/torque control according to the embodiment will be describedwith reference to FIGS. 6A and 6B. FIGS. 6A and 6B are diagramsillustrating voltage/torque control according to the embodiment.

FIG. 6A illustrates an example of a magnetic flux plane of a stator-sidecoordinate system. FIG. 6B illustrates an example of a magnetic fluxplane of a re-aligned coordinate system aligned with reference to therotor qds-axis magnetic flux estimated value λqdr_s_est(k+α). In FIGS.6A and 6B, ds axis is disposed downward in the drawing, and qs axis isdisposed rightward in the drawing. In the re-aligned coordinate systemillustrated in FIG. 6B, a direction of a vector (arrow) of the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1) is aligned withthe direction of the ds axis, and is illustrated as the rotor qds-axismagnetic flux estimated value λqdr_ras_est(k+α). Each magnetic fluxplane includes a torque line Te(k+2) derived from a physical model ofthe motor 2, a magnetic flux circle λc(k+2) representing the instructedintensity of the stator magnetic flux, and a range of the statormagnetic flux that can be output in the next cycle. The exampleillustrated in FIG. 6B is acquired by transforming a state of thestator-side coordinate system illustrated in FIG. 6A into a re-alignedcoordinate system.

First, FIG. 6A will be described.

Control variables used by the DB-DTFC include an air gap torque commandvalue Te_com(k+1) and a stator magnetic flux command valueλqds_s_com(k+2). A radius of the magnetic flux circle λc(k+2) is definedby a magnitude of the stator magnetic flux command valueλqds_s_com(k+2).

Here, in the power conversion unit 8, each of an upper arm and a lowerarm is composed of one switching device. Accordingly, the powerconversion unit 8 is composed of six switching devices. There are atotal of eight switching patterns of the six switching devices. Inaccordance with these eight switching patterns, a voltage vector thatcan be output from the power conversion unit 8, as illustrated in FIG.6A, has a hexagonal shape. Accordingly, a range of the output from thepower conversion unit 8 stays the inside of the hexagon. For example,the dqs-axis coordinate system is disposed so that its origin matchesthe center of the hexagon. Likewise, the re-aligned coordinate system ofFIG. 6B is also disposed so that its origin matches the center of thehexagon.

A torque line is a set of points representing a condition in which theamount of change in the torque is constant. A torque line Te(k+2)projected onto a magnetic flux plane of the stator qds-axis coordinatesystem is drawn as a straight line in which the points are connectedtogether. The torque line Te(k+2) defines a stator qds-axis magneticflux command value qds_s_com(k+2) for acquiring a desired torque in thenext control cycle. This torque line Te(k+2) is determined using an airgap torque Te_com(k+1), the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α), and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+α).

Here, for the sake of facilitating analysis, a magnetic flux plane ofthe re-aligned coordinate system illustrated in FIG. 6B will be used.Variables depending on the coordinate system are transformed through acoordinate transformation. For example, the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α) and the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α) of the stator magnetic flux coordinatesystem are respectively transformed into a rotor qds-axis magnetic fluxestimated value λqdr_ras_est(k+α) and a stator qds-axis magnetic fluxestimated value λqds_ras_est(k+α) of the re-aligned coordinate system.As illustrated in FIG. 6B, when the rotor qds-axis magnetic fluxestimated value λqdr_ras_est(k+α) is aligned in parallel with the dsaxis, the torque line Te(k+2) is parallel to the ds axis. This drawingcorresponds to second solution to be described later.

In order to achieve the desired torque, it is necessary to supply acertain stator qds-axis voltage command value Vqds_ras_com(k+1) in aperiod (the sampling period t_(s)) of one control cycle using the powerconversion device 3. That the pointing end (tip) of the stator qds-axisvoltage command value Vqds_ras_com(k+1) in a period is on the torqueline Te(k+2) means the desired torque is achieved after a samplingperiod. The summation of the stator qds-axis magnetic flux estimatedvalue λqds_ras_est(k+1) and the stator qds-axis voltage command valueVqds_ras_com(k+1) in a sampling period is the new resultant statorqds-axis magnetic flux λqds_ras(k+2) after a sampling period.

As described above, the instructed intensity of the stator magnetic fluxis defined as a magnetic flux circle λc(k+2). An intersection betweenthe torque line Te(k+2) and the magnetic flux circle λc(k+2) is a pointrepresenting a required torque. There are two intersections between thetorque line Te(k+2) and the magnetic flux circle λc(k+2). Out of these,a point disposed on the inside of the hexagon that can be reached fromthe origin through control is extracted as a point representing therequired torque. The DB-DTFC calculation unit 14 may determine thestator qds-axis voltage command value Vqds_ras_com(k+1)×t_(s) such thatthe pointing end (tip) of the arrow (vector), of which starting point(tail) is disposed at the origin, representing the stator qds-axisvoltage command value Vqds_ras_com(k+1) reaches the intersectiondescribed above.

The DB-DTFC calculation unit 14 will be described with reference to FIG.7. FIG. 7 is a block diagram illustrating the DB-DTFC calculation unit14 according to the embodiment.

The DB-DTFC calculation unit 14, for example, includes a torque lineprocessing unit 141, a magnetic flux circle processing unit 142, voltageconversion units 143 and 144, and an inverse RAS coordinatetransformation unit 145.

The torque line processing unit 141 calculates a voltage time productcommand value by performing a calculation process for specifying atorque line for the air gap torque command value Te_com(k+1). Thevoltage time product command value Vqds_ras_com(k+1)×t_(s) is an exampleof the voltage time product command value.

For example, the torque line processing unit 141 receives inputvariables including the air gap torque command value Te_com(k+1), thestator qds-axis magnetic flux estimated value λqds_s_est(k+1), the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1), the statorqds-axis magnetic flux estimated value λqds_s_est(k+α), the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+α), and the rotorangular velocity estimated value ωr_est(k+1), and calculates a voltagetime product command value Vqds_ras_com(k+1)×t_(s) corresponding to theair gap torque command value Te_com(k+1).

The magnetic flux circle processing unit 142 performs a calculationprocess for determining a voltage×time product defining a magnitude anda direction of a magnetic flux of the motor 2 using the re-alignedcoordinate system.

For example, the magnetic flux circle processing unit 142 receives inputvariables including the stator qds-axis magnetic flux estimated valueλqds_s_est(k+α), the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), the stator qds-axis magnetic flux command valueλqds_s_com(k+2), and the voltage time product command valueVqds_ras_com(k+1)×t_(s), and derives a desired voltage time productcommand value Vds_ras_com(k+1)×t_(s) on the basis of the magnetic fluxcircle λc on the magnetic flux plane of the re-aligned coordinate systemand the torque line Te described above.

The voltage conversion unit 143 calculates at least a q-axis componentof the voltage time product command value Vqs_ras_com(k+1) by dividing,by the sampling period t_(s), the voltage time product command valueVqds_ras_com(k+1)×t_(s) calculated by the torque line processing unit141.

The voltage conversion unit 144 calculates the voltage time productcommand value Vds_ras_com(k+1) of the d-axis component by dividing, bythe sampling period t_(s), the voltage time product command valueVqds_ras_com(k+1)×t_(s) calculated by the magnetic flux circleprocessing unit 142.

The inverse RAS coordinate transformation unit 145 calculates the statorqds-axis voltage command value Vqds_s_com(k+1) by performing an inverseRAS transformation for the voltage time product command valueVqs_ras_com(k+1) calculated by the voltage conversion unit 143 and thevoltage time product command value Vds_ras_com(k+1) calculated by thevoltage conversion unit 144 on the basis of the rotor qds-axis magneticflux estimated value λqdr_s_est(k+α).

For example, the RAS transformation is defined using the followingEquation (21).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 21} \right\rbrack & \; \\\begin{matrix}{\lambda_{dr}^{ras} = \sqrt{\lambda_{qr}^{s\; 2} + \lambda_{dr}^{s\; 2}}} \\{\lambda_{qr}^{ras} = 0} \\{\lambda_{ds}^{ras} = {{{\lambda_{qs}^{s} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}} + {\lambda_{ds}^{s} \times \frac{\lambda_{dr}^{s}}{\lambda_{dr}^{ras}}}} = \frac{{\lambda_{qs}^{s}\lambda_{qr}^{s}} + {\lambda_{ds}^{s}\lambda_{dr}^{s}}}{\lambda_{dr}^{ras}}}} \\{\lambda_{qs}^{ras} = {{{{- \lambda_{ds}^{s}} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}} + {\lambda_{qs}^{s} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}}} = \frac{{{- \lambda_{ds}^{s}}\lambda_{qr}^{s}} + {\lambda_{qs}^{s}\lambda_{dr}^{s}}}{\lambda_{dr}^{ras}}}}\end{matrix} & (21)\end{matrix}$

For example, an inverse RAS transformation corresponding to thedescription presented above is defined using the following Equation(22).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 22} \right\rbrack & \; \\\begin{matrix}{\lambda_{ds}^{s} = {{{{- \lambda_{qs}^{ras}} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}} + {\lambda_{ds}^{ras} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}}} = \frac{{{- \lambda_{qs}^{ras}}\lambda_{qr}^{s}} + {\lambda_{ds}^{ras}\lambda_{dr}^{s}}}{\lambda_{dr}^{ras}}}} \\{\lambda_{qs}^{s} = {{{\lambda_{ds}^{ras} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}} + {\lambda_{qs}^{ras} \times \frac{\lambda_{qr}^{s}}{\lambda_{dr}^{ras}}}} = \frac{{\lambda_{ds}^{ras}\lambda_{qr}^{s}} + {\lambda_{qs}^{ras}\lambda_{qr}^{s}}}{\lambda_{dr}^{ras}}}}\end{matrix} & (22)\end{matrix}$

A more specific example of each unit of the DB-DTFC calculation unit 14will now be described.

The torque line processing unit 141 includes, for example, a calculationblock 1410, a calculation block 1411A, a calculation block 1411B, acalculation block 1412, a calculation block 1413, a calculation block1414, a calculation block 1415, a calculation block 1416, a calculationblock 1417, a calculation block 1418, and a calculation block 1419.

The calculation block 1410 is an example of an estimated torquecalculation unit. For example, the calculation block 1410 receives inputvariables including the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1), and calculates an air gap torque estimated valueTe_est(k+1) (a torque estimated value). The calculation block 1410calculates an inner product of two vectors, i.e., the stator qds-axismagnetic flux estimated value λqds_s_est(k+1) and the rotor qds-axismagnetic flux estimated value λqdr_s_est(k+1), and multiplies the innerproduct by a predetermined coefficient K, thereby calculating an air gaptorque estimated value Te_est(k+1). The process described above isrepresented in Equation (23).[Math. 23]{circumflex over (T)} _(e)(k+1)=({circumflex over (λ)}_(qs)^(s)(k+1){circumflex over (λ)}_(dr) ^(s)(k+1)−{circumflex over (λ)}_(ds)^(s)(k+1){circumflex over (λ)}_(qr) ^(s)(k+1))×K   (23)

The calculation block 1412 multiplies the air gap torque estimated valueTe_est(k+1) by a coefficient represented in the following Equation (24).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 24} \right\rbrack & \; \\\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right) & (24)\end{matrix}$

The calculation block 1411A is a subtractor. The calculation block 1411Asubtracts the air gap torque estimated value Te_est(k+1) from the airgap torque command value Te_com(k+1) to acquire a difference thereof asΔTe_est(k+1). The calculation described above is represented in Equation(25).[Math. 25]Δ{circumflex over (T)} _(e)(k+1)=T* _(e)(k+1)−{circumflex over (T)}_(e)(k+1)   (25)

The calculation block 1411B is an adder. The calculation block 1411Badds ΔTe_est(k+1) calculated by the calculation block 1411A and theresult calculated by the calculation block 1412.

The calculation block 1413 multiplies a result calculated by the adder1411B by the following Equation (26).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 26} \right\rbrack & \; \\\frac{4\sigma\; L_{s}L_{r}}{3{PL}_{m}{{\hat{\lambda}}_{dr}^{s}\left( {k + 1} \right)}} & (26)\end{matrix}$

The calculation block 1414 normalizes a result calculated by thecalculation block 1413 on the basis of the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α). For example, the calculation block 1414divides a result acquired by the calculation block 1413 by the resultcalculated by the calculation block 1415. The output of the calculationblock 1414 represents a torque reference in the re-aligned coordinatesystem.

The calculation block 1415 calculates an absolute value (norm) of therotor qds-axis magnetic flux estimated value λqdr_ras_est(k+α) on thebasis of the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α). By using the result calculated by the calculation block1415, the re-aligned coordinate system can be made to match thestator-side coordinate system.

The calculation block 1416 executes an RAS coordinate transformation ofthe stator qds-axis magnetic flux estimated value λqds_s_est(k+α) withreference to the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α).

The calculation block 1417 multiplies the rotor angular velocityestimated value ωr_est(k+1) by the sampling period ts. The calculationblock 1418 calculates an inner product of a vector represented by aresult calculated by the calculation block 1416 and a vector representedby a result calculated by the calculation block 1417.

The calculation block 1419 adds the result calculated by the calculationblock 1414 and the result calculated by the calculation block 1418 toyield a voltage time product command value Vqs_ras_com(k+1)×t₀corresponding to the air gap torque command value Te_com(k+1).

The calculation block 1411A, the calculation block 1411B, thecalculation block 1412, the calculation block 1413, the calculationblock 1414, the calculation block 1415, the calculation block 1416, thecalculation block 1417, the calculation block 1418, and the calculationblock 1419 described above are an example of a torque line processingunit.

As described above, the torque line processing unit 141 calculates avoltage×time product corresponding to the product of the average voltageand the control period on the basis of at least the air gap torquecommand value Te_com(k+1) (the torque command for the motor 2), the airgap torque estimated value Te_est(k+1) (the torque estimated value forthe motor 2), the stator qds-axis magnetic flux estimated valueλqds_s_est(k+α) (the second estimated value of the stator magnetic fluxof the motor 2), the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α) (the second estimated value of the rotor magnetic fluxof the motor 2), and the rotation velocity ωr of the rotor magnetic fluxof the motor 2.

The magnetic flux circle processing unit 142 includes, for example, acalculation block 1420, a calculation block 1421, a calculation block1422, a calculation block 1423, a calculation block 1424, a calculationblock 1425, a calculation block 1426, a calculation block 1427, acalculation block 1428, a calculation block 1429, and a calculationblock 1430.

The calculation block 1420 calculates a stator qds-axis magnetic fluxestimated value λqds_ras_est(k+1) by executing a RAS coordinatetransformation of the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) with reference to the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α).

The calculation block 1422 calculates a stator qds-axis currentestimated value Iqds_ras_est(k+α) by executing a RAS coordinatetransformation of the stator qds-axis current estimated valueIqds_s_est(k+α) with reference to the rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+α). The calculation block 1423 multipliesthe stator qds-axis magnetic flux estimated value λqds_ras_est(k+α) by(t_(s)×Rs). The calculation block 1421 is a subtractor. The calculationblock 1421 subtracts the product of the stator qds-axis currentestimated value Iqds_ras_est(k+α) and (t_(s)×Rs) from the rotor qds-axismagnetic flux estimated value λqdr_ras_est(k+1). In accordance with aresult of the calculation described above, the center of the magneticflux circle is corrected using the product of the magnitude of thestator qds-axis current estimated value Iqds_ras_est(k+α) and the statorresistance Rs, and accordingly, the accuracy of the position of themagnetic flux circle λc is improved. When the product of the magnitudeof the stator qds-axis current estimated value Iqds_ras_est(k+α) and thestator resistance Rs is sufficiently small with respect to the radius ofthe magnetic flux circle, the process of correcting the center of themagnetic flux circle may be omitted.

By using the process of each of the calculation block 1420 to thecalculation block 1423 described above, a voltage×time productcorresponding to a voltage drop according to the stator qds-axis currentestimated value Iqds_ras_est(k+α) flowing through the stator resistanceRs is subtracted from the stator qds-axis magnetic flux estimated valueλqds_ras_est(k+1) in re-aligned coordinates, whereby a stator qds-axismagnetic flux estimated value λqds_ras_est(k+1)β reflecting theinfluence of the stator resistance Rs is acquired. The stator qds-axismagnetic flux estimated value λqds_ras_est(k+1)β depends on themagnitude of the stator qds-axis current estimated valueIqds_ras_est(k+α).

The calculation block 1424 adds the result calculated by the torque lineprocessing unit 141 and the result calculated by the calculation block1421.

The calculation block 1425 includes a D-axis coordinate calculationunit. For example, the calculation block 1425 calculates a position ofan intersection between the torque line and the magnetic flux circle λcon the basis of the stator qds-axis magnetic flux command valueλqds_s_com(k+2) and the result calculated by the calculation block 1424as the D-axis coordinate calculation unit. Details thereof will bedescribed later.

The calculation block 1426 subtracts the result calculated by thecalculation block 1421 from the result calculated by the calculationblock 1425.

The magnetic flux circle processing unit 142 described above determinesthe radius of the magnetic flux circle on the basis of a command valueof the rotor magnetic flux of the motor 2. The magnetic flux circleprocessing unit 142 determines the position of the center of themagnetic flux circle λc on the basis of the stator qds-axis magneticflux estimated value λqds_ras_est(k+α) (the second estimated value) andthe stator qds-axis magnetic flux estimated value λqds_ras_est(k+1) (thefirst estimated value) of the stator magnetic flux of the motor 2. Themagnetic flux circle processing unit 142 adjusts the position of thecenter of the magnetic flux circle λc on the basis of the statorqds-axis current estimated value Iqds_ras_est(k+α) (the second estimatedvalue) and the resistance value Rs of the stator winding. In this way,the influence of a voltage drop occurring in accordance with theresistance value Rs of the stator winding is alleviated using the statorqds-axis current estimated value Iqds_ras_est(k+α) (the second estimatedvalue), whereby the accuracy of the control can be improved.

Details of the DB-DTFC will be described.

First, equivalent equations of an induction motor in an arbitrarycoordinate system are represented in Equation (27-1) to Equation (27-4).[Math. 27]V _(qds) =R _(b) i _(qds)+(p+jω _(e))λ_(qds)   (27-1)0=R _(r) i _(qdr)+[p+j(ω_(e)−ω_(r)))]λ_(qdr)   (27-2)λ_(qds) =L _(s) i _(qds) +L _(m) i _(qdr)   (27-3)λ_(qdr) =L _(m) i _(qds) +L _(r) i _(qdr)   (27-4)

By solving Equation (27-3) and Equation (27-4) described above, a statorcurrent and a rotor current are acquired using the following Equation(28-1) and Equation (28-2).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 28} \right\rbrack & \; \\{i_{qds} = {{{\frac{L_{r}}{{L_{s}L_{r}} - L_{m}^{2}}\lambda_{qds}} - {\frac{L_{m}}{{L_{s}L_{r}} - L_{m}^{2}}\lambda_{qdr}}} = {{\frac{1}{\sigma\; L_{s}}\lambda_{qds}} - {\frac{L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qdr}}}}} & \left( {28\text{-}1} \right) \\{i_{qdr} = {{{{- \frac{L_{m}}{{L_{s}L_{r}} - L_{m}^{2}}}\lambda_{qds}} + {\frac{L_{r}}{{L_{s}L_{r}} - L_{m}^{2}}\lambda_{qdr}}} = {{{- \frac{L_{m}}{\sigma\; L_{s}L_{r}}}\lambda_{qds}} + {\frac{1}{\sigma\; L_{s}}\lambda_{qdr}}}}} & \left( {28\text{-}2} \right)\end{matrix}$

Here, a leakage coefficient u is defined as represented in the followingEquation (29).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 29} \right\rbrack & \; \\{\sigma = \frac{{L_{s}L_{r}} - L_{m}^{2}}{L_{s}L_{r}}} & (29)\end{matrix}$

By substituting Equation (28-1), Equation (28-2), and Equation (29) intoEquation (27-1) and Equation (27-2), the following Equation (30-1) andEquation (30-2) are acquired.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 30} \right\rbrack & \; \\{V_{qds} = {{\frac{R_{s}}{\sigma\; L_{s}}\lambda_{qds}} - {\frac{R_{s}L_{m}}{\sigma\; L_{s}}\lambda_{qdr}} + {\left( {p + {j\;\omega_{e}}} \right)\lambda_{qds}}}} & \left( {30\text{-}1} \right) \\{0 = {{{- \frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}}\lambda_{qds}} + {\frac{R_{r}}{\sigma\; L_{s}}\lambda_{qdr}} + {\left\lbrack {p + {j\left( {\omega_{e} - \omega_{r}} \right)}} \right\rbrack\lambda_{qdr}}}} & \left( {30\text{-}2} \right)\end{matrix}$

The following Equation (31-1) and Equation (31-2) are calculated bytransforming the following Equation (30-1) and Equation (30-2).

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 31} \right\rbrack & \; \\{{p\;\lambda_{qds}} = {V_{qds} - {\left( {\frac{R_{s}}{\sigma\; L_{s}} + {j\;\omega_{e}}} \right)\lambda_{qds}} + {\frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qdr}}}} & \left( {31\text{-}1} \right) \\{{p\;\lambda_{qdr}} = {{{- \left\lbrack {\frac{R_{r}}{\sigma\; L_{r}} + {j\left( {\omega_{e} - \omega_{r}} \right)}} \right\rbrack}\lambda_{qdr}} + {\frac{R_{r}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qds}}}} & \left( {31\text{-}2} \right)\end{matrix}$

In Equation (31-1) and Equation (31-2) described above, by substituting“0” into ωe using the characteristics of the rotary coordinate systemthrough the RAS coordinate transformation, the following Equation (32-1)and Equation (32-2) are calculated.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 32} \right\rbrack & \; \\{{p\;\lambda_{qds}^{s}} = {V_{qds}^{s} - {\frac{R_{s}}{\sigma\; L_{s}}\lambda_{qds}^{s}} + {\frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qdr}^{s}}}} & \left( {32\text{-}1} \right) \\{{p\;\lambda_{qdr}^{s}} = {{\frac{R_{r}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qds}^{s}} - {\left( {\frac{R_{r}}{\sigma\; L_{r}} + {j\;\omega_{r}}} \right)\lambda_{qdr}^{s}}}} & \left( {32\text{-}2} \right)\end{matrix}$

The following Equation (33) is an equation used for calculating thetorque of the motor 2.

$\begin{matrix}\left\lbrack {{Math}.\mspace{14mu} 33} \right\rbrack & \; \\{T_{e} = {{\frac{3{PL}_{m}}{4}\left( {{i_{qs}i_{dr}} - {i_{ds}i_{qr}}} \right)} = {\frac{3P}{4}\frac{L_{m}}{\sigma\; L_{s}L_{r}}\left( {{\lambda_{qs}\lambda_{dr}} + {\lambda_{ds}\lambda_{qr}}} \right)}}} & (33)\end{matrix}$

By differentiating Equation (33) described above, the following Equation(34) representing a change rate of the torque is calculated.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 34} \right\rbrack & \; \\{{\overset{.}{T}}_{e} = {\frac{3P}{4}\frac{L_{m}}{\sigma\; L_{s}L_{r}}\left( {{{\overset{.}{\lambda}}_{qs}\lambda_{dr}} + {\lambda_{qs}{\overset{.}{\lambda}}_{dr}} - {{\overset{.}{\lambda}}_{ds}\lambda_{qr}} - {\lambda_{ds}{\overset{.}{\lambda}}_{qr}}} \right)}} & (34)\end{matrix}$

Equation (31-1) and Equation (31-2) described above can be transformedinto Equation (35-1) to Equation (35-4) denoted as scalars.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 35} \right\rbrack & \; \\{{\overset{.}{\lambda}}_{qs} = {V_{qs} - {\frac{R_{s}}{\sigma\; L_{s}}\lambda_{qs}} - {\omega_{e}\lambda_{ds}} + {\frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qr}}}} & \left( {35\text{-}1} \right) \\{{\overset{.}{\lambda}}_{ds} = {V_{qs} - {\frac{R_{s}}{\sigma\; L_{s}}\lambda_{ds}} - {\omega_{e}\lambda_{qs}} + {\frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{dr}}}} & \left( {35\text{-}2} \right) \\{{\overset{.}{\lambda}}_{qr} = {{{- \frac{R_{r}}{\sigma\; L_{r}}}\lambda_{qr}} - {\left( {\omega_{e} - \omega_{r}} \right)\lambda_{dr}} + {\frac{R_{r}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qs}}}} & \left( {35\text{-}3} \right) \\{{\overset{.}{\lambda}}_{dr} = {{{- \frac{R_{r}}{\sigma\; L_{r}}}\lambda_{dr}} - {\left( {\omega_{e} - \omega_{r}} \right)\lambda_{qr}} + {\frac{R_{r}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{ds}}}} & \left( {35\text{-}4} \right)\end{matrix}$

By substituting Equation (35-1) to Equation (35-4) into Equation (34)described above, the following Equation (36) can be represented.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 36} \right\rbrack} & \; \\{{\overset{.}{T}}_{e} = {{\frac{3{PL}_{m}}{4\sigma\; L_{s}L_{r}}\left\lbrack {{v_{qs}\lambda_{dr}} - {v_{ds}\lambda_{qr}} - {\omega_{r}\left( {{\lambda_{qs}\lambda_{qr}} + {\lambda_{ds}\lambda_{dr}}} \right)}} \right\rbrack} - {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right)T_{e}}}} & (36)\end{matrix}$

When Equation (36) described in a continuous time system is transformedinto the following Equation (37) in a discrete time system.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 37} \right\rbrack} & \; \\{{{{\overset{.}{T}}_{e}\left( {k + 1} \right)} \approx \frac{\Delta\; T_{e}}{t_{s}}} = {\frac{{T_{e}^{*}\left( {k + 1} \right)} - {{\hat{T}}_{e}\left( {k + 1} \right)}}{t_{s}} = {{\frac{3{PL}_{m}}{4\sigma\; L_{s}L_{r}}\left\lbrack {{{V_{qs}\left( {k + 1} \right)}{\lambda_{dr}\left( {k + \alpha} \right)}} - {{V_{ds}\left( {k + 1} \right)}{\lambda_{qr}\left( {k + \alpha} \right)}} - {{\omega_{r}\left( {k + 1} \right)}\left( {{{\lambda_{qs}\left( {k + \alpha} \right)}{\lambda_{qr}\left( {k + \alpha} \right)}} + {{\lambda_{ds}\left( {k + \alpha} \right)}{\lambda_{dr}\left( {k + \alpha} \right)}}} \right)}} \right\rbrack} - {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right){{\hat{T}}_{e}\left( {k + 1} \right)}}}}} & (37)\end{matrix}$

By transforming Equation (37) representing the change rate of thetorque, Equation (38) is calculated.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 38} \right\rbrack} & \; \\{{{V_{qs}\left( {k + 1} \right)}t_{s}} = {{\frac{{\hat{\lambda}}_{qr}\left( {k + \alpha} \right)}{{\hat{\lambda}}_{dr}\left( {k + \alpha} \right)}{V_{ds}\left( {k + 1} \right)}t_{s}} + {\frac{4\sigma\; L_{s}L_{r}}{3{PL}_{m}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}\left\lbrack {{\Delta\;{{\hat{T}}_{e}\left( {k + 1} \right)}} + {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right){{\hat{T}}_{e}\left( {k + 1} \right)}t_{s}}} \right\rbrack} + {{\omega_{r}\left( {k + 1} \right)}t_{s}\frac{{{{\hat{\lambda}}_{qs}\left( {k + \alpha} \right)}{{\hat{\lambda}}_{qr}\left( {k + \alpha} \right)}} + {{{\hat{\lambda}}_{ds}\left( {k + \alpha} \right)}{{\hat{\lambda}}_{dr}\left( {k + \alpha} \right)}}}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}}} & (38)\end{matrix}$

Equation (38) described above is an equation representing a torque line.By defining a variable m as represented in Equation (39-2) and defininga variable b as represented in Equation (39-3), Equation (38) can besimplified as Equation (39-1).

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 39} \right\rbrack} & \; \\{\mspace{79mu}{{{V_{qs}\left( {k + 1} \right)}t_{s}} = {{m \times {V_{ds}\left( {k + 1} \right)}t_{s}} + b}}} & \left( {39\text{-}1} \right) \\{\mspace{79mu}{m = \frac{{\hat{\lambda}}_{qr}\left( {k + 1} \right)}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}} & \left( {39\text{-}2} \right) \\{b = {{\frac{4\sigma\; L_{s}L_{r}}{3{PL}_{m}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}\left\lbrack {{\Delta\;{{\hat{T}}_{e}\left( {k + 1} \right)}} + {\left( {\frac{R_{s}}{\sigma\; L_{s}} + \frac{R_{r}}{\sigma\; L_{r}}} \right){{\hat{T}}_{e}\left( {k + 1} \right)}t_{s}}} \right\rbrack} + {\omega_{r}t_{s}\frac{{{{\hat{\lambda}}_{qs}\left( {k + 1} \right)}{{\hat{\lambda}}_{qr}\left( {k + 1} \right)}} + {{{\hat{\lambda}}_{ds}\left( {k + 1} \right)}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}}{{\hat{\lambda}}_{dr}\left( {k + 1} \right)}}}} & \left( {39\text{-}3} \right)\end{matrix}$

As described above, in the case of second solution, m corresponds to theslope with respect to the ds axis in FIG. 6B, and thus m=0. When m is“0”, a qs-axis component Vqs_ras_com(k+1) of the voltage command valuein the re-aligned coordinate system is calculated using the followingEquation (40).[Math. 40]V _(qs)(k+1)=b/t _(s)   (40)

Next, the process relating to the magnetic flux circle λc will bedescribed.

A differential value of the stator qds-axis magnetic flux λqds can berepresented as in the following Equation (41).[Math. 41]pλ _(qds) =V _(qds) −R _(s) i _(qds) −jω _(c)λ_(qds)   (41)

Equation (41) described above is transformed into Equation (42) of adiscrete time system.[Math. 42]{circumflex over (λ)}_(qds) ^(s)(k+2)=(V _(qs) ^(s)(k+1)−R _(s) î _(qs)^(s)(k+α))t _(s)+{circumflex over (λ)}_(qds) ^(s)(k+1)   (42)

The Equation (42) is separated into qs-axis component and ds-axiscomponent as shown in Equation (43).[Math. 43]|{circumflex over (λ)}_(qds) ^(s)(k+2)|²=(r _(qs))²+(r _(ds))²r _(qs)=(V _(qs) ^(s)(k+1)−R _(s) î _(qs) ^(s)(k+α))×t _(S)+{circumflexover (λ)}_(qs) ^(s)(k+1)r _(ds)=(V _(ds) ^(s)(k+1)−R _(s) î _(ds) ^(s)(k+α))×t _(S)+{circumflexover (λ)}_(ds) ^(s)(k+1)   (43)

In Equation (43) described above, the qs-axis component Vqs_s_com(k+1)of the voltage command value in the stator stationary coordinate systemcan be acquired as a known variable using Equation (42). By solvingEquation (43) described above, the ds-axis component Vds_s_com(k+1) ofthe voltage command value is calculated.

In the description presented above, although the stator qds-axismagnetic flux estimated value λqds_s_est(k+2) has been used, it can besubstituted for by the stator qds-axis magnetic flux command valueλqds_s_com(k+1) as described above.

In the description presented above, although the stator stationarycoordinate system has been illustrated as an example, the re-alignedcoordinate system can also be used in a similar manner. The qs-axiscomponent Vqs_ras_com(k+1) of the voltage command value can be acquiredas a known variable using an equation of the re-aligned coordinatesystem transformed from Equation (42). By solving the equation of there-aligned coordinate system transformed from Equation (43), the ds-axiscomponent Vds_ras_com(k+1) of the voltage command value can be acquired.

According to the embodiment described above, the current/magnetic fluxestimation unit 20 generates the stator qds-axis current estimated valueIqds_s_est(k+1), the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) at a higher degree of estimation accuracy. The averagecorrection unit 30 generates the stator qds-axis current estimated valueIqds_s_est(k+α), the stator qds-axis magnetic flux estimated valueλqds_s_est(k+α), and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+α) on the basis of the stator qds-axis current estimatedvalue Iqds_s_est(k+1), the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1), and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1). The DB-DTFC calculation unit 14 generates the statordqs-axis voltage command value Vdqs_ras_com(k+1) using at least thestator qds-axis magnetic flux estimated value λqds_s_est(k+α) and therotor qds-axis magnetic flux estimated value λqdr_s_est(k+α). In thisway, the control device 10 can improve the accuracy of control of themotor 2.

In some cases, the accuracy of control may be improved by compensatingthe amount of delay in the calculation executed by the current observer21. This will be described with reference to FIG. 22. FIG. 22 is adiagram illustrating advantages of a compensation for the amount ofdelay in calculation executed by the current observer 21 according tothe embodiment. In graphs illustrated in FIG. 22, a stator magnetic fluxλqds_s corresponding to elapse of time (seconds) is represented by athick solid line, a stator current Iqds_s is represented by a thin solidline, and a rotor magnetic flux λqdr_s is represented by double lines.Each line represents actual changes from a time point k to a time point(k+1). Each line later than the time point (k+1), including a time point(k+2), represents the behavior of the motor 2 estimated at the timepoint (k+1).

The current observer 21 calculates a moving-time average value save ofthe stator magnetic flux that is an average of the stator magnetic fluxmeasured value λqds_s_est(k) and a stator magnetic flux measured valueλqds_s_est(k+1) at the time point (k+1). A value represented by a pointG11 in FIG. 22 is a moving-time average value save of the statormagnetic flux. As illustrated in FIG. 22, the point G11 is away from theline of the stator magnetic flux λ. A point G12 on the line of thestator magnetic flux λ representing the same value as the point G11 isin the past from the time point (k+1) by (t_(s)/2).

The current observer 21 calculates a stator current estimated valueIqds_s_est(k+1) using the moving average value λave of the statormagnetic flux as described above. A value represented by a point G21 inFIG. 22 is a stator current estimated value Iqds_s_est(k+1). However,like the case of the stator magnetic flux λqds_s, the point G21 in FIG.22 is away from the line of the stator current Iqds_s at the time point(k+1). A point G22 on the line of the stator current I representing thesame magnitude as the point G21 is in the past from the time point (k+1)by (t_(s)/2).

In the process of calculating the rotor magnetic flux estimated valueλqdr_s_est(k+1), the magnetic flux observer 22 uses the stator currentestimated value Iqds_s_est(k+1) described above. The rotor magnetic fluxestimated value λqdr_s_est(k+1) includes a component of the statorcurrent estimated value Iqds_s_est(k+1). In the rotor magnetic fluxestimated value λqdr_s_est(k+1), a delay of about (t_(s)/2)substantially occurs as described above.

The delay described above may be included in values of the statevariables of the motor 2 calculated by the current observer 21 and themagnetic flux observer 22.

A state from the time point k to the time point (k+2) changes inaccordance with elapse of time. However, when variation in the value ofthe synchronous angular frequency ωe is deemed to be a constant and itis assumed that the tendency of change in the state from the time pointk to the time point (k+2) is continuous, the change in the state can bepredicted. The average correction unit 30 calculates a state quantity ata time point (k+α) using the value of the state variable of the timepoint (k+1) using the synchronous angular frequency ωe. In this way, theDB-DTFC calculation unit 14 can use the predicted value of the statequantity of the time point (k+α) as a value of the state variable forthe time point (k+1). For example, when the time point (k+α) is adjustedto be in the future from the time point (k+1) by (t_(s)/2), the delayoccurring in the process of calculation executed by the current observer21 matches the period of time from the time point (k+1) to the timepoint (k+α), thus offsetting the influences thereof.

For example, when about a time of (t_(s)/2) elapses from the time pointat the point G22 illustrated in the graph in the middle of FIG. 22, itwill be the time point (k+1). During that period, the state at the timepoint of the point G22 changes along the line of the stator currentIqds_s and is detected as a stator current measured valueIqds_s_det(k+1). Since much noise is included in the stator currentmeasured value Iqds_s_det(k+1), the stator current estimated valueIqds_s_est(k+1) calculated by the current observer 21 is used forcontrol instead of the stator current measured value.

Thus, the average correction unit 30 estimates the stator currentestimated value Iqds_s_est(k+α) using the synchronous angular frequencyωe on the basis of the stator current estimated value Iqds_s_est(k+1)corresponding to the point G21, whereby acquiring the state quantity atthe position of the point G23 of the time point (k+α). The statequantity at the position of the point G23 is the stator currentestimated value Iqds_s_est(k+α). By using this value as the statequantity at the time point (k+1), the stator current estimated valueIqds_s_est(k+α) can be used as if it is the state quantity at the timepoint (k+1). A point G24 represents the stator current estimated valueIqds_s_est(k+α) used as a state quantity of the time point (k+1).

The above explanation using the stator current Iqds_s can also besimilarly applied to cases using the stator magnetic flux λqds_s or therotor magnetic flux λqdr_s described above. A point G11, a point G13,and a point G14 in the stator magnetic flux λqds_s respectivelycorrespond to the point G21, the point G23, and the point G24 of thestator current Iqds_s. A point G31, a point G33, and a point G34 in therotor magnetic flux λqdr_s respectively correspond to the point G21, thepoint G23, and the point G24 of the stator current Iqds_s.

The average correction unit 30, as described above, corrects a value ofthe variable representing each state quantity using a predeterminedtranslation rule to offset a delay occurring due to a calculationprocess executed by the current/magnetic flux estimation unit 20.Accordingly, the phase of the stator qdr-axis magnetic flux estimatedvalue λqdr_s_est used by the DB-DTFC calculation unit 14 as a referenceis adjusted to offset the delay described above, and accordingly, theaccuracy of the phase of the stator dqs-axis voltage command valueVdqs_ras_com(k+1) generated by the DB-DTFC calculation unit 14 isimproved.

According to the embodiment, the accuracy of control executed by theDB-DTFC can be improved by reducing the influence of voltage dropoccurring due to an impedance component of the stator resistance Rs.

When the influence caused by the voltage drop due to the impedancecomponent of the stator resistance Rs is small, the second term inEquation (42) described above may be omitted.

First Modification of First Embodiment

A first modification of the embodiment will be described with referenceto the drawings.

In the first embodiment, a case relating to the second solution in theDB-DTFC control has been described. In this modification, instead ofthis, a case relating to the first solution in the DB-DTFC control willbe described.

FIG. 8 is a diagram illustrating the first solution in the DB-DTFCcontrol according to the embodiment. FIG. 8 illustrates an example of amagnetic flux plane of a magnetic flux coordinate system. The magneticflux coordinate system illustrated in FIG. 8 is a re-aligned coordinatesystem aligned with reference to the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α). ds axis is disposed downward in thedrawing, and qs axis is disposed rightward in the drawing.

FIG. 8 is different from FIG. 6B in that the stator qds-axis magneticflux estimated value λqds_s_est(k+α) is aligned with the ds axis. Inthis case, a torque line Te(k+2) is not parallel to the ds axis.

Thus, in the first solution, an intersection between a magnetic fluxcircle and a torque line is derived using a method for acquiring asolution of a quadratic equation by simultaneously solving an equationrepresenting the magnetic flux circle and an equation representing thetorque line.

For the simplification of the description, Equation (43) representingthe magnetic flux circle λc is transformed into Equation (44). ThisEquation (44) is simplified by disregarding the influence of the statorresistance Rs.[Math. 44]|{circumflex over (λ)}_(qds) ^(ras)(k+2)|²=(V _(qs) ^(ras)(k+1)t_(s))²+(V _(ds) ^(ras)(k+1)t _(s)+{circumflex over (λ)}_(ds)^(ras)(k+1))²   (44)

By simultaneously solving the equations of the magnetic flux circuit andthe torque line, the equations can be transformed into the followingEquation (45). By solving this, a qs-axis component Vqs_ras_com(k+1) ofthe voltage command value can be acquired from Equation (46), and ads-axis component Vds_ras_com(k+1) of the voltage command value can beacquired from Equation (45-3).

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 45} \right\rbrack} & \; \\{{{\left( {m^{2} + 1} \right)\left\lbrack {{V_{ds}^{ras}\left( {k + 1} \right)}t_{s}} \right\rbrack}^{2} + {\left( {{2{mb}} + {2{\lambda_{ds}^{ras}\left( {k + 1} \right)}}} \right)\left\lbrack {{V_{ds}^{ras}\left( {k + 1} \right)}t_{s}} \right\rbrack} + b^{2} + \left( {\lambda_{ds}^{ras}\left( {k + 1} \right)} \right)^{2} - {{\lambda_{qds}^{*}\left( {k + 1} \right)}}^{2}} = 0} & (45) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 46} \right\rbrack} & \; \\{{{V_{ds}^{{ras}*}\left( {k + 1} \right)}t_{s}} = {\frac{- \left( {{mb} + {\lambda_{ds}^{ras}\left( {k + 1} \right)}} \right)}{m^{2} + 1} + \frac{\sqrt{\begin{matrix}{\left( {{mb} + {\lambda_{ds}^{ras}\left( {r + 1} \right)}} \right)^{2} - \left( {m^{2} + 1} \right)} \\\left\lbrack {b^{2} - {{\lambda_{qds}^{*}\left( {k + 1} \right)}}^{2} + \left( {\lambda_{ds}^{ras}\left( {k + 1} \right)} \right)^{2}} \right\rbrack\end{matrix}}}{m^{2} + 1}}} & (46) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 47} \right\rbrack} & \; \\{\mspace{79mu}{{{V_{qs}^{ras}\left( {k + 1} \right)}t_{s}} = {{m \times {V_{ds}^{ras}\left( {k + 1} \right)}t_{s}} + b}}} & (47)\end{matrix}$

According to the modification described above, the re-aligned coordinatesystem aligned with reference to the stator qds-axis magnetic fluxestimated value λqds_s_est(k+α) is used. This modification can alsoachieve the effects similar to those of the embodiment described above.

Second Modification of First Embodiment

A second modification of the embodiment will be described with referenceto the drawings.

In the first embodiment, a case has been described in which the latchinterface is used for the stator magnetic flux in the process executedby the current observer 21. In this modification, instead of this, acase will be described in which a ramp interface is used for the statormagnetic flux. This ramp interface functions as a first-order holdingfunction.

A modification of the current observer 21 will be described withreference to FIG. 9.

FIG. 9 is a block diagram illustrating a current observer 21A accordingto the second modification of the first embodiment. Instead of thecurrent observer 21 illustrated in FIG. 1 described above, the currentobserver 21A calculates an estimated value of a current flowing throughthe stator winding of the motor on the basis of a drive quantity commandvalue and the estimated value of the stator magnetic flux calculated asdescribed above. For example, instead of the calculation block 211, thecalculation block 212, the calculation block 213, and the calculationblock 218 of the current observer 21, the current observer 21A includesa calculation block 212A, a calculation block 213A, a calculation block213B, a calculation block 218A, a calculation block 218B, and acalculation block 219.

The calculation block 212A calculates a voltage V2 by multiplying astator qds-axis magnetic flux estimated value λqds_s_est(k+1) by atransfer function having variables including the rotor resistance Rr,the rotor winding inductance Lr, and the rotor velocity ωr representedin Equation (48).

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 48} \right\rbrack & \; \\{\frac{R_{r}}{L_{r}} - {j\;\omega_{r}}} & (48)\end{matrix}$

The calculation block 219 divides a result of calculation executed bythe calculation block 212A by (χσLs) and outputs a quotient thereof.

The calculation block 218B includes a ramp interface. The calculationblock 218B performs first-order holding of the result of the calculationexecuted by the calculation block 219 and calculates a voltagecorrection value Vcomp1 which is the result of the first-order holding.The calculation block 218B smooths the stator qds-axis magnetic fluxestimated value λqds_s_est using time history data of the statorqds-axis magnetic flux estimated value λqds_s_est.

For example, the calculation block 218B calculates a voltage correctionvalue Vcomp1 by multiplying a transfer function represented in Equation(49) having variables A1, B0, and B1 by the result of the calculationexecuted by the calculation block 219.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 49} \right\rbrack & \; \\\frac{B_{0} + {B_{1}z^{- 1}}}{1 - {A_{1}z^{- 1}}} & (49)\end{matrix}$

In Equation (49) described above, A1, B0, and B1 are defined using thefollowing Equation (50). Here, χ is an arbitrary variable.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 50} \right\rbrack & \; \\{{A_{1} = e^{{- \chi}\;{ts}}}{B_{0} = {1 - \frac{1}{\chi\;{ts}} + {\frac{1}{\chi\;{ts}}e^{{- \chi}\;{ts}}}}}{B_{1} = {\frac{1}{\chi\;{ts}} - {\frac{1}{\chi\;{ts}}e^{{- \chi}\;{ts}}} - e^{{- \chi}\;{ts}}}}} & (50)\end{matrix}$

As described above, the calculation block 218B includes at least a rampcalculation unit that performs first-order holding of a value based onan estimated value of the rotor magnetic flux.

The calculation block 212A, the calculation block 219, and thecalculation block 218B described above are an example of a currentestimated value calculation unit. The calculation block 212A, thecalculation block 219, and the calculation block 218B calculate anestimated value (a first estimated value) of a current flowing throughthe stator winding of the motor 2 on the basis of an estimated value ofthe stator magnetic flux of the motor 2 calculated by the magnetic fluxobserver 22. The calculation block 218B performs first-order holding ofa value based on the estimated value of the stator magnetic flux in thecalculation process of calculating a first estimated value of thecurrent flowing through the stator winding of the motor 2 from theestimated value of the stator magnetic flux of the motor 2.

The calculation block 213A is an adder. The calculation block 213A addsthe stator qds-axis voltage command value Vqds_s_com, the result ofcalculation executed by the calculation block 215, and the result ofcalculation executed by the calculation block 216, and outputs theresult of addition as the voltage sum value Vqds_sum1. The voltage sumvalue Vqds_sum1 does not include a component based on the statorqds-axis magnetic flux estimated value λqds_s_est(k+1). The calculationblock 217 divides, by (χσLs), the voltage sum value Vqds_sum1 notincluding the component of the stator qds-axis magnetic flux estimatedvalue λqds_s_est(k+1). It should be noted that the component based onthe stator qds-axis magnetic flux estimated value is added in thecalculation block 213B in a later stage.

The calculation block 218A may be the same as the calculation block 218described above. The calculation block 218A performs a predeterminedcalculation process on the basis of a result of the calculation executedby the calculation block 217 and outputs a result thereof (Iqds_sum2).Iqds_sum2 does not include a component based on the stator qds-axismagnetic flux estimated value λqds_s_est(k+1).

The calculation block 213B is an adder. The calculation block 213B addsa result of the calculation executed by the calculation block 218A and aresult of the calculation executed by the calculation block 218B andoutputs the result of addition as a stator qds-axis current estimatedvalue Iqds_s_est(k+1). The calculation block 210 according to themodification maintains the stator qds-axis current estimated valueIqds_s_est(k+1) that is a result of the calculation executed by thecalculation block 213B.

The description of the configuration of the current observer 21A will becontinued.

In a transformation from a continuous system to a discrete system, thecurrent observer 21A applies a latch interface to a voltage input andapplies a ramp interface to a magnetic flux interface.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 51} \right\rbrack & \; \\{{pi}_{qds}^{s} = {\frac{1}{\sigma\; L_{s}}\left\lbrack {V_{qds}^{s} + {\left( {\frac{Rr}{Lr} - {j\;\omega_{r}}} \right)\lambda_{qds}^{s}} - {R_{eq}i_{qds}^{s}} + {j\;\omega_{r}\sigma\; L_{s}i_{qds}^{s}}} \right\rbrack}} & (51)\end{matrix}$

In Equation (51) described above, Equation (52) is defined.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 52} \right\rbrack & \; \\{V_{2}^{s} = {\left( {\frac{Rr}{Lr} - {j\;\omega_{r}}} \right)\lambda_{qds}^{s}}} & (52)\end{matrix}$

Since Equation (51) is based on linear system, the following Equation(53) is acquired by applying superposition principle.

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 53} \right\rbrack & \; \\{{\frac{i_{qds}^{s}}{V_{qds}^{s}} = {\frac{1}{\sigma\; L_{s}}\frac{1}{s + \chi}}}{\frac{i_{qds}^{s}}{V_{2}^{s}} = {\frac{1}{\sigma\; L_{s}}\frac{1}{s + \chi}}}} & (53)\end{matrix}$

χ in Equation (53) described above is defined as represented in thefollowing Equation (54).

$\begin{matrix}\left\lbrack {{Math}.\mspace{11mu} 54} \right\rbrack & \; \\{\chi = {\frac{R_{eq}}{\sigma\; L_{s}} - {j\;\omega_{r}}}} & (54)\end{matrix}$

As described above, by applying the latch interface to a stator voltage,applying the ramp interface to a stator magnetic flux, and transformingEquation (54) into an equation of a discrete time system, Equation (55)and Equation (56) are acquired. Z[ ] represents Z transformation.

$\begin{matrix}{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 55} \right\rbrack} & \; \\{\mspace{79mu}{\frac{i_{qds}^{s}(z)}{V_{qds}^{s}(z)} = {{\frac{1}{\sigma\; L_{s}}\left( {1 - z^{- 1}} \right){Z\left\lbrack \frac{1}{s\left( {s + \chi} \right)} \right\rbrack}} = {\frac{1}{\sigma\; L_{s}}\frac{1}{\chi}\frac{\left( {1 - e^{{- \chi}\; t_{s}}} \right)z^{- 1}}{1 - {e^{{- \chi}\; t_{s}}z^{- 1}}}}}}} & (55) \\{\mspace{79mu}\left\lbrack {{Math}.\mspace{11mu} 56} \right\rbrack} & \; \\{\frac{i_{qds}^{s}(z)}{V_{2}^{s}(z)} = {{\frac{1}{\sigma\; L_{s}}\frac{\left( {1 - z^{- 1}} \right){Z\left\lbrack \frac{1}{s^{2}\left( {x + \chi} \right)} \right\rbrack}}{\left( {1 - z^{- 1}} \right){Z\left\lbrack \frac{1}{s^{2}} \right\rbrack}}} = {\frac{1}{\sigma\; L_{s}}\frac{1}{\chi}\frac{\begin{matrix}{\left( {1 - \frac{1}{\chi\; t_{s}} + {\frac{1}{\chi\; t_{s}}e^{{- \chi}\; t_{s}}}} \right) +} \\{\left( {\frac{1}{\chi\; t_{s}} - {\frac{1}{\chi\; t_{s}}e^{{- \chi}\; t_{s}}} - e^{{- \chi}\; t_{s}}} \right)z^{- 1}}\end{matrix}}{1 - {e^{{- \chi}\; t_{s}}z^{- 1}}}}}} & (56)\end{matrix}$

Here, the following Equation (57) is acquired by defining A₁, B₀, and B₁as described above and simplifying Equation (55) and Equation (56).[Math. 57]î _(qds) ^(s)(k+1)=K×[(1−A ₁)V _(qds) ^(s)(k+1)+B ₀ ×V′ ₂(k+1)+B ₁ ×V′₂(k)]+A ₁ ×i′ _(qds)(k)e ^(−χt) ^(s)   (57)

According to the modification described above, even when the currentobserver 21A receives a magnetic flux estimated value using the latchinterface, effects similar to those according to the embodiment areacquired.

Second Embodiment

A motor drive system 1A according to a second embodiment will bedescribed with a focus on a DB-DTFC calculation unit 14A. In the firstembodiment, for example, the accuracy of estimation of a current and amagnetic flux is improved using the observer, and a value of a statevariable (state quantity) in the DB-DTFC control is corrected throughaverage correction. In this second embodiment, DB-DTFC control in whichthe accuracy of estimation of a current and a magnetic flux is improvedby the observer but an effect resulting from correcting a value of thestate variable through the average correction described above is notacquired.

FIG. 10 is a block diagram illustrating the motor drive system 1Aaccording to the second embodiment.

A control device 10A of the motor drive system 1A is different from thecontrol device 10 of the motor drive system 1 described above in that,e.g., the DB-DTFC calculation unit 14A is included instead of theDB-DTFC calculation unit 14, the average correction unit 30 is notincluded, and the slip frequency estimation unit 18 and the adder unit19 can be omitted. FIG. 10 does not illustrate the slip frequencyestimation unit 18 and the adder unit 19.

The DB-DTFC calculation unit 14A receives input variables including anair gap torque command value Te_com(k+1), a rotor qds-axis magnetic fluxestimated value λqdr_s_est(k+1), a stator qds-axis magnetic fluxestimated value λqds_s_est(k+1), and a stator qds-axis magnetic fluxcommand value λqds_s_com(k+2) and calculates a stator qds-axis voltagecommand value Vqds_s_com(k+1) on the basis of the input variablesdescribed above. The stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) are supplied from a current/magnetic flux estimationunit 20. The stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) is an example of an estimated value of a stator magneticflux of a motor 2. The rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) is an example of an estimated value of a rotor magneticflux of the motor 2.

While the DB-DTFC calculation unit 14 described above performs RAScoordinate transformation and inverse RAS coordinate transformation withreference to the rotor gds-axis magnetic flux estimated valueλqdr_s_est(k+α), the DB-DTFC calculation unit 14A performs RAScoordinate transformation and inverse RAS coordinate transformation withreference to the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1).

FIG. 11 is a block diagram illustrating the DB-DTFC calculation unit 14Aaccording to the embodiment.

Instead of the torque line processing unit 141, the magnetic flux circleprocessing unit 142, and the inverse RAS coordinate transformation unit145 of the DB-DTFC calculation unit 14 described above, the DB-DTFCcalculation unit 14A, includes a torque line processing unit 141A, amagnetic flux circle processing unit 142A, and an inverse RAS coordinatetransformation unit 145A.

The torque line processing unit 141A includes a calculation block 1415Aand a calculation block 1416A instead of the calculation block 1415 andthe calculation block 1416 of the torque line processing unit 141.

The calculation block 1415A calculates a magnitude (norm) of the statorqds-axis magnetic flux estimated value λqds_s_est(k+1).

The calculation block 1416A performs a RAS coordinate transformation ofa stator qds-axis magnetic flux estimated value λqds_s_est(k+1) withreference to the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) to yield a stator qds-axis magnetic flux estimated valueλqds_ras_est(k+1). As a result of the calculation described above, thestator qs-axis magnetic flux estimated value λqs_ras_est(k+1) is zero,and the stator ds-axis magnetic flux estimated value λds_ras_est(k+1) isthe magnitude of the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1).

The magnetic flux circle processing unit 142A is different from themagnetic flux circle processing unit 142 in that the magnetic fluxcircle processing unit 142A does not include the calculation block 1421,the calculation block 1422, and the calculation block 1423, and includesa calculation block 1420A in place of the calculation block 1420.

The calculation block 1420A has the same configuration as that of thecalculation block 1416A described above. The calculation block 1420Asupplies the result of calculation, i.e., the stator qds-axis magneticflux estimated value λqds_ras_est(k+1), to the calculation block 1424and the calculation block 1426 disposed in a later stage.

As described above, in this second embodiment, the function of a part ofthe motor drive system 1 according to the first embodiment is degraded,and the configuration can be simplified. The DB-DTFC calculation unit14A can acquire a desired solution using the technique of firstsolution.

Third Embodiment

A motor drive system 1B according to a third embodiment will bedescribed.

In the embodiments described above, for example, the deadbeat-directtorque control (DB-DTFC) system is used for generating a voltage commandvalue. This third embodiment uses a field oriented control system isused. Hereinafter, the field oriented control system will be referred toas an FOC system.

In the FOC system, a current component generating a torque (a rotationforce) and a current component generating a magnetic flux aredecomposed, and each current component is independently controlled as adirect current amount.

An example of FOC system includes a direct field oriented control (DFOC)system. In the DFOC system, a magnetic flux vector is directly estimatedby a magnetic flux sensor or a magnetic flux observer and is controlled.

FIG. 12 is a block diagram illustrating the motor drive system 1Baccording to the embodiment. The motor drive system 1B illustrated inthis drawing includes a control device 10B based on the FOC systeminstead of the control device 10 of the motor drive system 1 describedabove. The motor 2 does not have a sensor detecting a position or avelocity.

Examples of FOC systems include an indirect field oriented control(IFOC) system and a direct field oriented control (DFOC) system. TheIFOC system uses indirect vector control (also referred to as slipfrequency-type vector control) controlling slip of an induction motorwithout estimation or detection of a magnetic flux. The DFOC system usesdirect vector control controlling slip of the motor 2 on the basis of aresult of estimation or detection of a magnetic field.

A DFOC calculation unit 40 illustrated in FIG. 12 is an example based onthe DFOC system described above. The control device 10B has the DFOCcalculation unit 40 in place of the DB-DTFC calculation unit 14, a sixthcoordinate transformation unit 15A in place of the first coordinatetransformation unit 15, a velocity/phase estimation unit 13A in place ofthe velocity/phase estimation unit 13, and a slip frequency estimationunit 18A in place of the slip frequency estimation unit 18.

The control device 10B includes a seventh coordinate transformation unit17A and an eighth coordinate transformation unit 17B. A v-phase statorcurrent Ivs(k) and a w-phase stator current Iws(k) are supplied to theseventh coordinate transformation unit 17A. Three-phase stator voltagecommand values Vus_com(k+1), Vvs_com(k+1), and Vws_com(k+1) that areoutputs of the sixth coordinate transformation unit 15A are supplied tothe eighth coordinate transformation unit 17B.

A torque command value Te_com(k+1) calculated by a motion controller 12is supplied to the DFOC calculation unit 40. A stator magnetic fluxcommand value λs_com of a rated stator magnetic flux is supplied to theDFOC calculation unit 40 as a magnetic flux command.

The velocity/phase estimation unit 13A calculates, for example, a rotorangular velocity estimated value ωrm_est(k+1), a rotor angle estimatedvalue θr_est(k+1), and a stator power source phase estimated valueθe_est(k+1) on the basis of the torque command value Te_com(k+1)calculated by the motion controller 12, the slip angle frequencyestimated value ωsl_est(k+1) calculated by the slip frequency estimationunit 18A, and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1).

The velocity/phase estimation unit 13A, for example, includes a positiontracking observer. The position tracking observer described above usesthe stator qds-axis magnetic flux estimated value λqds_s_est(k+1) as areference signal for extracting a phase.

For example, the velocity/phase estimation unit 13A extracts afundamental component (a synchronous angle θe) on the basis of thestator qds-axis magnetic flux estimated value λqds_s_est(k+1) andgenerates a rotor angle θr by correcting the fundamental component onthe basis of the slip angle frequency estimated value ωsl_est(k+1). Thevelocity/phase estimation unit 13A compares the rotor angle θr with therotor angle estimated value θr_est(k) and generates a rotor angleestimated value θr_est(k+1) adjusted to have a corresponding phase. Thevelocity/phase estimation unit 13A generates a rotor angular velocityestimated value ωrm_est(k+1) in accordance with the rotor angleestimated value θr_est(k+1).

The slip frequency estimation unit 18A calculates, for example, a slipangle frequency estimated value ωsl_est(k+1) relating to slip of themotor 2 on the basis of the torque command value Te_com(k+1) calculatedby the motion controller 12 and the stator qds-axis magnetic fluxestimated value λqds_s_est(k+1) calculated by the magnetic flux observer22.

The velocity/phase estimation unit 13A outputs the stator power sourcephase estimated value θe_est(k+1) as a reference signal to the seventhcoordinate transformation unit 17A and the sixth coordinatetransformation unit 15A. The seventh coordinate transformation unit 17Atransforms input signals into signals of γ component and component thatare orthogonal to each other in the rotary coordinate system on thebasis of the reference signal.

By appropriately selecting the phase of the reference signal, theseventh coordinate transformation unit 17A can obtain, as γ component, acomponent having the same phase as the reference phase, and obtain, as δcomponent, a component orthogonal to the reference phase. The seventhcoordinate transformation unit 17A supplies, to the DFOC calculationunit 40, a γ-axis stator current Iγ(k+1) and a δ-axis stator currentIδ(k+1) of which the coordinates have been transformed.

Oppositely to the above, the sixth coordinate transformation unit 15Atransforms a voltage command value Vγ_com(k+1) of the γ component and avoltage command value Vδ_com(k+1) of the δ component that are outputsfrom the DFOC calculation unit 40 into three-phase stator voltagecommand values Vus_com(k+1), Vvs_com(k+1), and Vws_com(k+1) of thestator coordinate system. The result of the calculation executed by thesixth coordinate transformation unit 15A is supplied to the PWMcontroller 16 and the eighth coordinate transformation unit 17B.

The eighth coordinate transformation unit 17B transforms the three-phasestator voltage command values Vus_com(k+1), Vvs_com(k+1), andVws_com(k+1) into a voltage command value Vqds_s_com(k+1) of a two-axiscomponent of the dqs axis. The value transformed by the eighthcoordinate transformation unit 17B is supplied to the current/magneticflux estimation unit 20.

The rotor qds-axis magnetic flux estimated value λqdr_s_est(k+1) and thestator qds-axis magnetic flux estimated value λqds_s_est(k+1) estimatedby the current/magnetic flux estimation unit 20 are supplied to the DFOCcalculation unit 40.

The DFOC calculation unit 40 generates a stator current command valueIγ_com(k+1) of the γ component and a stator current command valueIδ_com(k+1) of the δ component on the basis of the supplied signals, andgenerates a stator voltage command value Vγ_com(k+1) and a statorvoltage command value Vδ_com(k+1) such that the γ-axis stator currentIγ(k+1) and the δ-axis stator current Iδ(k+1) follow the command values.

The voltage command value Vγ_com(k+1) of the γ component and the voltagecommand value Vδ_com(k+1) of the δ component are supplied to the sixthcoordinate transformation unit 15A and are given to the power conversiondevice 3 further through the PWM controller 16 as a gate pulse GP.

In the third embodiment described above, although the control system isdifferent from that of the first and second embodiments, thecurrent/magnetic flux estimation unit 20 is disposed in the controldevice, and the rotor qds-axis magnetic flux estimated valueλqdr_s_est(k+1) and the stator qds-axis magnetic flux estimated valueλqds_s_est(k+1) supplied from the current/magnetic flux estimation unit20 are used for control, which are similar to the second embodiment. Inaccordance with the current/magnetic flux estimation unit 20, an effectof improving the accuracy of estimation of the rotor qds-axis magneticflux estimated value λqdr_s_est(k+1) and the stator qds-axis magneticflux estimated value λqds_s_est(k+1) is acquired.

The control device 10B can use the accuracy of estimation of the rotorqds-axis magnetic flux estimated value λqdr_s_est(k+1) and the statorqds-axis magnetic flux estimated value λqds_s_est(k+1) estimated by thecurrent/magnetic flux estimation unit 20. Therefore, the accuracy ofcontrol using the FOC system can be further improved.

The control system according to the embodiment is the FOC. In the caseof the FOC, a feedback signal of a current (a current FBK) is used forcontrol using the DFOC calculation unit 40. For example, when theaccuracy of estimation of the stator current Idqs_est using the currentobserver 21 is ideal, an estimated current of the time point (k+1) (thestator current Idqs_est) can be used for control instead of the currentFBK based on a measured value. In this case, the DFOC calculation unit40 can perform zero-lag current control.

In the motor 2 according to the embodiment, a physical velocity sensoris not disposed. In the case of the embodiment, velocity sensorlesscontrol using an estimated magnetic flux using the magnetic fluxobserver 22 is performed. In the case of the velocity sensorlesscontrol, the improvement of a current estimation accuracy contributes tothe improvement of a magnetic flux estimation accuracy and furthercontributes to the improvement of an accuracy of the estimated phase andvelocity, and the control accuracy is improved. A control accuracy inthe case of the FOC is an accuracy of current control and velocitycontrol.

The velocity sensorless control can also be applied to the DB-DTFCdescribed above. When the DB-DTFC performs velocity sensorless control,a control accuracy means an accuracy in velocity control, magnetic fluxcontrol, and torque control.

Modification of Third Embodiment

In the third embodiment, for example, the average correction unit 30 isnot included, but the configuration is not limited thereto. The averagecorrection unit 30 may be included. In this case, an accuracy ofestimation of a magnetic flux and a current can be improved more greatlythan the third embodiment. When the sampling period ts is relativelylong, the average correction unit 30 is expected to be able to improvean accuracy of control.

Control Device According to Embodiment

A control device 10 according to an embodiment will be described. FIG.13 is a block diagram illustrating the control device 10 according tothe embodiment. The control device 10 includes a processing circuit 100.The processing circuit 100 illustrated in FIG. 13 includes a CPU 101, astorage unit 102, and a drive unit 103. The CPU 101, the storage unit102, and the drive unit 103 are interconnected through a bus. Theprocessing circuit 100 is an example of the control device 10. The CPU101 includes a processor executing a desired process in accordance witha software program. The storage unit 102 includes a semiconductormemory. The drive unit 103 generates a control signal of the powerconversion device 3 in accordance with control of the CPU 101. In theembodiment, processes executed by the CPU 101 and the drive unit 103will be collectively described simply as a process of the control device10. For example, the control device 10 controls the power conversiondevice 3 on the basis of detection results acquired by the currentdetectors 9 a, 9 b, and the like.

EXAMPLE

An evaluation result according to an example will be described withreference to FIGS. 14 to 18.

FIGS. 14 and 15 are diagrams illustrating evaluation results of a motordrive system 1 according to an examples. FIG. 14 illustrates anevaluation result (First example) when the current observer 21illustrated in FIG. 2 is applied to the control device 10 illustrated inthe first embodiment. FIG. 15 illustrates an evaluation result (Secondexample) when the current observer 21 illustrated in FIG. 2 is appliedto the control device 10A illustrated in the first embodiment.

On the other hand, FIG. 16 is a diagram illustrating an evaluationresult of a motor drive system according to a comparative example. Inthe comparative example illustrated in FIG. 16, a current observeraccording to the comparative example is applied in the control device10A illustrated in the first embodiment. The current observer accordingto the comparative example is different from the current observer 21 inthe following points. The first difference is that a rotor qds-axismagnetic flux estimated value λqdr_s_est(k+1) generated by the magneticflux observer is used as an input variable. The second difference isthat moving-time averaging for a magnetic flux input is not performed,and a ramp interface is not included. As described above, the currentobserver according to the comparative example does not perform a processof smoothing a magnetic flux input.

The evaluation results illustrated in each of FIGS. 14 to 16 areevaluation results when a switching frequency fs is set to 240 hertz(Hz), and a step response test of a torque is performed when the motor 2is driven at a rated velocity. Hereinafter, these will be comparativelydescribed.

From the evaluation result of the first example illustrated in FIG. 14,it can be read that a command value of a torque, a measured value, andan estimated value coincide with each other in a desired range inaccordance with the control using the control device 10.

From the evaluation result of the second example illustrated in FIG. 15,it can be read that a measured value and an estimated value aredifferent from a command value of a torque in the control using thecontrol device 10A. Here, while the measured value and the estimatedvalue are different from the torque command value described above, themeasured value and the estimated value respond to a stepped change inthe torque. The above-described evaluation result acquired by thecontrol device 10A can be considered to be evaluating a value of anactual torque.

In the evaluation result according to the comparative exampleillustrated in FIG. 16, a command value of a torque, a measured value,and an estimated value are independent from each other. Although it wasconfirmed that stepped changes of the values are synchronized with eachother, correlations among the values like those found in the first andsecond examples were not found.

Therefore, under the above test conditions, it was confirmed that it maybe difficult to apply the comparative example, and the configuration ofthe first example exhibits effective results.

A result of a step response test of a torque will be described. FIG. 17is a diagram illustrating an evaluation result of a step response testof a torque. Graphs represented in FIG. 17 analytically illustrateresults of a step response test of a torque. The vertical axisrepresents the magnitude of torque error, the horizontal axis representsthe velocity of a rotor, and a depth represents the magnitude of atorque.

Step response tests of a torque were performed with two conditions,i.e., 2000 hertz (Hz) and 240 hertz (Hz) as the sampling frequency fs.As a state when the tests were executed, the motor 2 was driven in asteady state, and the torque and a load were changed in a predeterminedrange. Error between the torque estimated value and a measured value atthat time was represented in three-dimensional graphs. As predeterminedranges of the torque and the load, the torque was changed in a rangefrom zero to a rated load, and the velocity was changed in a range fromzero to a rated velocity.

In any of the case of 200 hertz (Hz) and 240 hertz (Hz), a result wasobtained such that the motor drive system 1 according to the embodimentachieves a lower error than that of the comparative example.Particularly, in the case of 240 hertz (Hz), the difference wassignificant. From the evaluation result of 240 hertz (Hz), it can beread that the motor drive system 1 can be controlled without increasingtorque error even when the velocity of the motor 2 is increased.

In order to verify the evaluation results described above, thecurrent/magnetic flux estimation unit 20 was evaluated.

FIG. 18 is a diagram illustrating evaluation results of thecurrent/magnetic flux estimation unit 20. The evaluation resultsillustrated in FIG. 18 include an evaluation result of thecurrent/magnetic flux estimation unit 20 and an evaluation result of thecurrent/magnetic flux estimation unit according to the comparativeexample. The evaluation result (A: SECOND EXAMPLE) of thecurrent/magnetic flux estimation unit 20 is an evaluation resultobtained with the control device 10A according to the second embodimenthaving the current observer 21 (FIG. 2). The evaluation result (B:COMPARATIVE EXAMPLE) of the current/magnetic flux estimation unit is anevaluation result obtained with the control device 10A having thecurrent observer according to the comparative example instead of thecurrent observer 21.

In FIG. 18, comparative results of the stator magnetic flux λqs_s andthe stator magnetic flux estimated value λqs_s_est are illustrated on inthe first graph from the top, comparative results of the stator currentIqs_s and the stator current estimated value Iqs_s_est are illustratedin the second graph from the top, and comparative results of the rotormagnetic flux λqr_s and the rotor magnetic flux estimated valueλqr_s_est are illustrated in the third graph from the top, and threecomparative results of the torque command value Te_com, the torquemeasured value Te_det, and the torque estimated value Te_est areillustrated in the graph at the bottom. The conditions of the tests wereas follows: the sampling frequency fs is set to 500 hertz (Hz), the loadof the motor 2 is zero, and a predetermined velocity limit is applied.The sampling frequency fs described above was set to 10 times thefundamental frequency.

Here, such evaluation results illustrated in FIG. 18 will becomparatively described.

In the case of the second example, it can be determined that, in any oneof the stator magnetic flux, the stator current, the rotor magneticflux, and the torque, a stepped quantization distortion appears in theestimated value, but the reproducibility of an input waveform is high,and an input waveform is followed.

On the other hand, in the case of the comparative example, thereproducibility of an input waveform is low for the stator current andthe rotor magnetic flux. Regarding the torque, a result was acquiredsuch that the torque measured value Te_det and the torque estimatedvalue Te_est are different from the torque command value Te_com. In thecase of the comparative example, there are values not following an inputwaveform.

According to at least one of the embodiments described above, thecontrol device 10 of the power conversion device 3 includes the DB-DTFCcalculation unit 14, the PWM controller 16, the magnetic flux observer22, the current observer 21, and the average correction unit 30.Accordingly, in the PWM controller 16, the power conversion device 3driving the motor 2 controls current flowing through the winding of themotor 2. The DB-DTFC calculation unit 14 calculates a drive quantitycommand value defining a drive quantity of the motor 2 and controls thepower conversion device 3 on the basis of the calculated drive quantitycommand value. The magnetic flux observer 22 calculates a firstestimated value of the stator magnetic flux of the motor 2 and a firstestimated value of the rotor magnetic flux of the motor 2 on the basisof at least the calculated drive quantity command value. The currentobserver 21 calculates a first estimated value of the current flowingthrough the stator winding of the motor 2 on the basis of at least thecalculated estimated value of the stator magnetic flux. The averagecorrection unit 30 includes the first transformation processing unit 31,the second transformation processing unit 32, and the thirdtransformation processing unit 33. The first transformation processingunit 31 calculates a second estimated value of the stator magnetic fluxof the motor 2 for the first estimated value of the stator magnetic fluxof the motor 2 in accordance with the predetermined transformation rulebased on the synchronous/excitation angular frequency ωe of the motor 2.The second transformation processing unit 32 calculates a secondestimated value of the rotor magnetic flux of the motor 2 for the firstestimated value of the rotor magnetic flux of the motor 2 in accordancewith the predetermined transformation rule based on thesynchronous/excitation angular frequency ωe of the motor 2. The thirdtransformation processing unit 33 calculates a second estimated value ofthe stator current of the motor 2 for the first estimated value of thestator current of the motor 2 in accordance with the predeterminedtransformation rule based on the synchronous/excitation angularfrequency ωe of the motor 2. The DB-DTFC calculation unit 14 calculatesa control quantity defining a drive quantity of the motor 2 on the basisof at least the torque command for the motor 2, the second estimatedvalue of the stator magnetic flux of the motor 2, the second estimatedvalue of the rotor magnetic flux of the motor 2, and the secondestimated value of the current flowing through the stator winding of themotor 2. Accordingly, an induction motor can be controlled with a highaccuracy.

While certain embodiments have been described, these embodiments havebeen presented by way of example only, and are not intended to limit thescope of the inventions. Indeed, the novel embodiments described hereinmay be embodied in a variety of other forms, furthermore, variousomissions, substitutions and changes in the form of the embodimentsdescribed herein may be made without departing from the spirit of theinventions. The accompanying claims and their equivalents are intendedto cover such forms or modifications as would fall within the scope andspirit of the inventions.

What is claimed is:
 1. A control device for a power conversion devicethat drives a motor comprising: a drive control unit that controls acurrent which the power conversion device causes to flow through astator winding of the motor; a drive quantity adjustment unit thatcalculates a drive quantity command value defining a drive quantity ofthe motor and controls the power conversion device on the basis of thecalculated drive quantity command value; a magnetic flux observer thatcalculates a first estimated value of a stator magnetic flux of themotor and a first estimated value of a rotor magnetic flux of the motoron the basis of at least the calculated drive quantity command value; acurrent observer that calculates a first estimated value of the currentflowing through the stator winding of the motor on the basis of at leastthe calculated first estimated value of the stator magnetic flux; and anaverage correction unit that includes a first transformation processingunit, a second transformation processing unit, and a thirdtransformation processing unit, wherein the first transformationprocessing unit calculates a second estimated value of the statormagnetic flux of the motor for the first estimated value of the statormagnetic flux of the motor in accordance with a first predeterminedtransformation rule based on an synchronous/excitation angular frequencyof the motor, wherein the second transformation processing unitcalculates a second estimated value of the rotor magnetic flux of themotor for the first estimated value of the rotor magnetic flux of themotor in accordance with a second predetermined transformation rulebased on the synchronous/excitation angular frequency of the motor,wherein the third transformation processing unit calculates a secondestimated value of a stator current of the motor for the first estimatedvalue of the stator current of the motor in accordance with a thirdpredetermined transformation rule based on the synchronous/excitationangular frequency of the motor, and wherein the drive quantityadjustment unit calculates a control quantity defining the drivequantity of the motor on the basis of at least a torque command for themotor, the second estimated value of the stator magnetic flux of themotor, the second estimated value of the rotor magnetic flux of themotor, and the second estimated value of the current flowing through thestator winding of the motor.
 2. The control device for the powerconversion device according to claim 1, wherein the drive quantityadjustment unit includes a magnetic flux circle calculation processingunit that determines a radius of a magnetic flux circle on the basis ofa command value of the rotor magnetic flux of the motor, determines aposition of the center of the magnetic flux circle on the basis of atleast the first estimated value of the stator magnetic flux of themotor, and adjusts the position of the center of the magnetic fluxcircle on the basis of the second estimated value of the current flowingthrough the stator winding of the motor and a resistance value of thestator winding.
 3. The control device for the power conversion deviceaccording to claim 1, wherein the drive quantity adjustment unitincludes a torque line processing unit that calculates a torqueestimated value of the motor on the basis of the first estimated valueof the stator magnetic flux of the motor and the first estimated valueof the rotor magnetic flux of the motor, wherein the torque lineprocessing unit that calculates a voltage time product corresponding toa product of an average voltage and a control period on the basis of atleast the torque command for the motor, the torque estimated value ofthe motor, the second estimated value of the stator magnetic flux of themotor, the second estimated value of the rotor magnetic flux of themotor, and a rotation speed of the rotor magnetic flux of the motor. 4.The control device for the power conversion device according to claim 1,wherein the average correction unit adjusts a correction coefficientaccording to any one of the first to third predetermined transformationrules in accordance with a magnitude of the synchronous/excitationangular frequency of the motor.
 5. The control device for the powerconversion device according to claim 3, wherein, in a calculation cycleallocated from the first time point as a start point to the second timepoint after the first time point of a discrete time system, the torqueline processing unit calculates the torque estimated value of the motoron the basis of the first estimated value of the stator magnetic flux ofthe motor and the first estimated value of the rotor magnetic flux ofthe motor at the first time point.
 6. The control device for the powerconversion device according to claim 5, wherein the torque lineprocessing unit calculates a voltage time product corresponding to aproduct of the average voltage and the control period on the basis of atleast the torque command for the motor acquired at the first time point,the torque estimated value of the motor at the first time point, thesecond estimated value of the stator magnetic flux of the motor, thesecond estimated value of the rotor magnetic flux of the motor, and therotation speed of the rotor magnetic flux of the motor at the first timepoint.
 7. The control device for the power conversion device accordingto claim 6, wherein the torque line processing unit calculates thevoltage time product on the basis of a result of a coordinatetransformation with reference to the second estimated value of thestator magnetic flux of the motor.
 8. The control device for the powerconversion device according to claim 2, wherein the magnetic flux circlecalculation processing unit determines a position of the center of themagnetic flux circle on the basis of a result of a coordinatetransformation with reference to the first estimated value of the statormagnetic flux of the motor.
 9. A motor drive system comprising: a powerconversion device; a control device for the power conversion deviceincluding: a drive control unit that controls a current which the powerconversion device causes to flow through a stator winding of the motor;a drive quantity adjustment unit that calculates a drive quantitycommand value defining a drive quantity of the motor and controls thepower conversion device on the basis of the calculated drive quantitycommand value; a magnetic flux observer that calculates a firstestimated value of a stator magnetic flux of the motor and a firstestimated value of a rotor magnetic flux of the motor on the basis of atleast the calculated drive quantity command value; a current observerthat calculates a first estimated value of the current flowing throughthe stator winding of the motor on the basis of at least the calculatedfirst estimated value of the stator magnetic flux; and an averagecorrection unit that includes a first transformation processing unit, asecond transformation processing unit, and a third transformationprocessing unit, wherein the first transformation processing unitcalculates a second estimated value of the stator magnetic flux of themotor for the first estimated value of the stator magnetic flux of themotor in accordance with a first predetermined transformation rule basedon an synchronous/excitation angular frequency of the motor, wherein thesecond transformation processing unit calculates a second estimatedvalue of the rotor magnetic flux of the motor for the first estimatedvalue of the rotor magnetic flux of the motor in accordance with asecond predetermined transformation rule based on thesynchronous/excitation angular frequency of the motor, wherein the thirdtransformation processing unit calculates a second estimated value of astator current of the motor for a first estimated value of the statorcurrent of the motor in accordance with a third predeterminedtransformation rule based on the synchronous/excitation angularfrequency of the motor, and wherein the drive quantity adjustment unitcalculates a control quantity defining the drive quantity of the motoron the basis of at least a torque command for the motor, the secondestimated value of the stator magnetic flux of the motor, the secondestimated value of the rotor magnetic flux of the motor, and the secondestimated value of the current flowing through the stator winding of themotor.
 10. A control device for a power conversion device comprising: adrive control unit that controls a current which the power conversiondevice causes to flow through a stator winding of the motor; a drivequantity adjustment unit that calculates a drive quantity command valuedefining a drive quantity of the motor and controls the power conversiondevice on the basis of the calculated drive quantity command value; amagnetic flux observer that calculates a first estimated value of astator magnetic flux of the motor and a first estimated value of a rotormagnetic flux of the motor on the basis of at least the calculated drivequantity command value; a current observer that calculates a firstestimated value of the current flowing through the stator winding of themotor on the basis of at least the calculated first estimated value ofthe stator magnetic flux; and an average correction unit that receivesthe first estimated value of the stator magnetic flux of the motor, thefirst estimated value of the rotor magnetic flux of the motor, and thefirst estimated value of the current flowing through the stator windingof the motor, and calculates a second estimated value of the statormagnetic flux of the motor, a second estimated value of the rotormagnetic flux of the motor, and a second estimated value of the currentflowing through the stator winding of the motor, by advancing phases ofthe first estimated value of the stator magnetic flux of the motor, thefirst estimated value of the rotor magnetic flux of the motor, and thefirst estimated value of the current flowing through the stator windingof the motor by a predetermined angle corresponding to asynchronous/excitation angular frequency of the motor.
 11. The controldevice for the power conversion device according to claim 10, whereinthe predetermined angle corresponding to the synchronous/excitationangular frequency of the motor is defined on the basis of a period inwhich the drive quantity adjustment unit calculates a drive quantitycommand value and the synchronous/excitation angular frequency.
 12. Amotor drive system comprising: a power conversion device; and a controldevice for the power conversion device including: a drive control unitthat controls a current which the power conversion device causes to flowthrough a stator winding of the motor; a drive quantity adjustment unitthat calculates a drive quantity command value defining a drive quantityof the motor and controls the power conversion device on the basis ofthe calculated drive quantity command value; a magnetic flux observerthat calculates a first estimated value of a stator magnetic flux of themotor and a first estimated value of a rotor magnetic flux of the motoron the basis of at least the calculated drive quantity command value; acurrent observer that calculates a first estimated value of the currentflowing through the stator winding of the motor on the basis of at leastthe calculated first estimated value of the stator magnetic flux; and anaverage correction unit that receives the first estimated value of thestator magnetic flux of the motor, the first estimated value of therotor magnetic flux of the motor, and the first estimated value of thecurrent flowing through the stator winding of the motor, and calculatesa second estimated value of the stator magnetic flux of the motor, asecond estimated value of the rotor magnetic flux of the motor, and asecond estimated value of the current flowing through the stator windingof the motor, by advancing phases thereof by a predetermined anglecorresponding to a synchronous/excitation angular frequency of themotor.
 13. A control device for a power conversion device comprising: adrive control unit that controls a current which the power conversiondevice causes to flow through a stator winding of the motor; a drivequantity adjustment unit that calculates a drive quantity command valuedefining a drive quantity of the motor through feedback control andcontrols the power conversion device on the basis of the calculateddrive quantity command value; a magnetic flux observer that calculates afirst estimated value of a stator magnetic flux of the motor and a firstestimated value of a rotor magnetic flux of the motor on the basis of atleast the calculated drive quantity command value; and an averagecorrection unit that adjusts the first estimated value of the statormagnetic flux of the motor and the first estimated value of the rotormagnetic flux of the motor on the basis of an estimated value of acontrol state of the motor and calculates feedback quantities of thefeedback control.
 14. A motor drive system comprising: a powerconversion device; and a control device for a power conversion devicecomprising: a drive control unit that controls a current which the powerconversion device causes to flow through a stator winding of the motor;a drive quantity adjustment unit that calculates a drive quantitycommand value defining a drive quantity of the motor through feedbackcontrol and controls the power conversion device on the basis of thecalculated drive quantity command value; a magnetic flux observer thatcalculates a first estimated value of a stator magnetic flux of themotor and a first estimated value of a rotor magnetic flux of the motoron the basis of at least the calculated drive quantity command value;and an average correction unit that adjusts the first estimated value ofthe stator magnetic flux of the motor and the first estimated value ofthe rotor magnetic flux of the motor on the basis of an estimated valueof a control state of the motor and calculates feedback quantities ofthe feedback control.